Rf transceiver with distributed filtering topology

ABSTRACT

This disclosure includes embodiments of a radio frequency (RF) transceiver having a distributed duplex filtering topology. The RF transceiver includes a power amplifier and a tunable RF duplexer. The tunable RF duplexer is configured to input an RF transmission input signal from the power amplifier, generate an RF transmission output signal that operates within an RF transmission band in response to the RF transmission input signal from the power amplifier, and simultaneously output the RF transmission output signal to an antenna and input an RF receive input signal that operates within an RF receive band from the antenna. The power amplifier includes a plurality of RF amplifier stages coupled in cascode and an RF filter coupled between a first one of the RF amplifier stages and a second one of the RF amplifier stages. Accordingly, the RF filter is configured to provide tuning within the RF receive band.

RELATED APPLICATIONS

This application claims the benefit of U.S. Provisional PatentApplication Ser. No. 61/595,795, filed on Feb. 7, 2012 and entitled“TUNABLE DUPLEXER,” the disclosure of which is hereby incorporatedherein by reference in its entirety.

The application is also related to U.S. patent application Ser. No.13/633,459, filed on Oct. 2, 2012 and entitled “TUNABLE DUPLEXERARCHITECTURE,” which claims the benefit of U.S. provisional patentapplication No. 61/542,939, filed on Oct. 4, 2011 and entitled “TUNABLEDUPLEXER ARCHITECTURE,” the disclosures of which are hereby incorporatedherein by reference in their entireties.

FIELD OF THE DISCLOSURE

This disclosure relates generally to radio frequency (RF) duplexers andduplexing methods related to RF front-end modules.

BACKGROUND

A duplexer is a device that facilitates bi-directional communication(i.e., simultaneous reception and emission) by a common antenna. Inorder to facilitate simultaneous reception and emission of signals overthe antenna, the duplexer needs to be designed for operation at bothreceive and transmission bands while providing adequate isolationbetween receive and transmission signals. To provide multi-mode andmulti-band operation, radio frequency (RF) duplexers often use multipleparallel duplexer components selected using an ever-growing number ofswitches. As such, this type of RF duplexer solution presentsever-increasing demands with regard to cost and size.

Accordingly, more adaptable RF duplexers are needed to providemulti-band/multi-mode operations that do not require increases in costand size.

SUMMARY

This disclosure includes embodiments of a radio frequency (RF)transceiver having a distributed duplex filtering topology. The RFtransceiver includes a power amplifier and a tunable RF duplexer. Thetunable RF duplexer is configured to input an RF transmission inputsignal from the power amplifier, generate an RF transmission outputsignal that operates within an RF transmission band in response to theRF transmission input signal from the power amplifier, andsimultaneously output the RF transmission output signal to an antennaand input an RF receive input signal that operates within an RF receiveband from the antenna. The power amplifier includes a plurality of RFamplifier stages coupled in cascode and an RF filter coupled between afirst one of the plurality of RF amplifier stages and a second one ofthe plurality of RF amplifier stages. Accordingly, the RF filter isconfigured to provide tuning within the RF receive band.

Those skilled in the art will appreciate the scope of the disclosure andrealize additional aspects thereof after reading the following detaileddescription in association with the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings incorporated in and forming a part of thisspecification illustrate several aspects of the disclosure, and togetherwith the description serve to explain the principles of the disclosure.

FIG. 1 illustrates one embodiment of a tunable radio frequency (RF)duplexer.

FIG. 2 illustrates exemplary procedures that may be implemented by thetunable RF duplexer shown in FIG. 1 so that the tunable RF duplexer maysimultaneously receive an RF receive input signal from an upstreamreceive chain while transmitting an RF transmission output signal to anantenna, and may receive an RF transmission input signal from adownstream transmission chain while providing an RF transmission outputsignal to the same antenna.

FIG. 2A is a frequency domain representation of one embodiment of an RFtransmission input signal that operates within an RF transmission bandand an RF receive output signal that operates within an RF receive band.

FIG. 2B is a frequency domain representation of a frequency response asS21 and S12 parameters provided by an RF filter circuit in the tunableRF duplexer shown in FIG. 1, wherein the frequency response defines apassband and a stopband.

FIG. 2C illustrates the frequency response shown in FIG. 2B after thepassband has been shifted to include the RF receive band shown in FIG.2A and the stopband has been shifted to include the RF transmission bandshown in FIG. 2A.

FIG. 2D illustrates the frequency response of the RF filter circuit asS11 and S22 parameters.

FIG. 2E is a frequency domain representation of one embodiment of afirst RF quadrature hybrid receive signal (QHRS) provided by splittingthe RF receive input signal, and a first RF quadrature hybridtransmission signal (QHTS) provided by splitting the RF transmissioninput signal.

FIG. 2F is a frequency domain representation of one embodiment of asecond RF QHRS provided by splitting the RF receive input signal and asecond RF QHTS provided by splitting the RF transmission input signal.

FIG. 2G is a frequency domain representation of one embodiment of the RFreceive output signal provided as a result of combining the first RFQHRS and the second RF QHRS and the RF transmission output signalprovided as a result of combining the first RF QHTS and the second RFQHTS.

FIG. 3A illustrates one embodiment of a tunable RF duplexer along with areceive signal flow.

FIG. 3B illustrates the same embodiment of the tunable RF duplexer alongwith a transmission signal flow.

FIG. 4 illustrates one embodiment of a frequency response of an RFfilter circuit in the tunable RF duplexer shown in FIGS. 3A and 3Bwherein the frequency response defines a stopband and a passband.

FIG. 5 illustrates another embodiment of a tunable RF duplexer, whereinthe tunable RF duplexer is designed for dual carrier aggregation.

FIG. 6A illustrates the tunable RF duplexer shown in FIG. 5 along with areceive signal flow.

FIG. 6B illustrates the tunable RF duplexer shown in FIG. 5 along with atransmission signal flow.

FIG. 7 illustrates one embodiment of a frequency response of an RFfilter circuit in the tunable RF duplexer shown in FIGS. 6A and 6Bwherein the frequency response defines a stopband and a passband.

FIG. 8A illustrates one embodiment of a tunable hybrid coupler that maybe provided in the tunable RF duplexer shown in FIGS. 6A and 6B.

FIG. 8B illustrates one embodiment of the tunable hybrid coupler shownin FIG. 8A formed in a semiconductor substrate.

FIG. 9A illustrates one embodiment of a tunable hybrid coupler that maybe provided in the tunable RF duplexer shown in FIGS. 6A and 6B.

FIG. 9B illustrates one embodiment of the tunable hybrid coupler shownin FIG. 9A formed in a semiconductor substrate.

FIGS. 10A-10E illustrate different embodiments of an RF filter that maybe used in the tunable RF duplexer shown in FIGS. 6A and 6B.

FIG. 11A illustrates an exemplary frequency response of any one of theRF filters shown in FIGS. 10A-10E.

FIG. 11B illustrates an RF communication channel having an RF receiveband and an RF transmission band separated by a duplex offset.

FIG. 12A illustrates one embodiment of a tunable RF duplexer that istunable to set the stopband within the RF transmission band and to setthe passband within the RF receive band whenever the RF receive band isat a higher frequency range than the RF transmission band.

FIG. 12B illustrates one embodiment of a tunable RF duplexer that istunable to set the stopband within the RF transmission band and to setthe passband within the RF receive band whenever the RF receive band isat a higher frequency range than the RF transmission band.

FIG. 12C illustrates one embodiment of a tunable RF duplexer that istunable to set the stopband within the RF transmission band and to setthe passband within the RF receive band whenever the RF receive band isat a higher frequency range than the RF transmission band.

FIG. 13A illustrates a transmission signal flow of the tunable RFduplexer shown in FIG. 12C.

FIG. 13B illustrates a receive signal flow of the tunable RF duplexershown in FIG. 12C.

FIG. 14 illustrates one embodiment of an RF transceiver having adistributed duplex filtering topology.

DETAILED DESCRIPTION

The embodiments set forth below represent the necessary information toenable those skilled in the art to practice the disclosure andillustrate the best mode of practicing the disclosure. Upon reading thefollowing description in light of the accompanying drawings, thoseskilled in the art will understand the concepts of the disclosure andwill recognize applications of these concepts not particularly addressedherein. It should be understood that these concepts and applicationsfall within the scope of the disclosure and the accompanying claims.

FIG. 1 illustrates an embodiment of a tunable radio frequency (RF)duplexer 10. An antenna 12 is operably associated with the tunable RFduplexer 10 and is capable of radiating RF receive signals and absorbingradiated RF transmission signals. In order to prevent out-of-band noiseand spurious emissions from distorting RF transmission and receivesignals, the tunable RF duplexer 10 provides isolation between RFreceive signals and RF transmission signals, as well as out-of-bandfiltering. Accordingly, the tunable RF duplexer 10 allows for an RFreceive input signal 14 to be intercepted by the antenna 12 whilesimultaneously emitting an RF transmission output signal 16 from theantenna 12.

In this embodiment, the tunable RF duplexer 10 includes a first hybridcoupler 18, a second hybrid coupler 20, an RF filter circuit 22, and atuning circuit 24. The RF filter circuit 22 includes an RF filter 23Aacross its bottom ports and an RF filter 23B across its top ports. Thus,the RF filter circuit 22 is a four-port network. However, the RF filter23A and the RF filter 23B are independent of one another and each of theRF filters 23A, 23B operates as a two-port network. Also, the RF filter23A and the RF filter 23B may be identical, and thus may each have thesame individual frequency response.

The first hybrid coupler 18 is operable to receive an RF transmissioninput signal 26 from upstream RF circuitry. For example, the RFtransmission input signal 26 is received from a power amplifier 28upstream from the tunable RF duplexer 10. The RF transmission outputsignal 16 is provided by the first hybrid coupler 18 at the antenna 12.An RF receive output signal 30 is transmitted by the second hybridcoupler 20 to a low noise amplifier (LNA) 31. Prior to the first hybridcoupler 18 receiving the RF transmission input signal 26, the RFtransmission input signal 26 is filtered by a second RF filter circuit32 to remove spurious emissions. As explained in further detail below,the first hybrid coupler 18, the RF filter circuit 22, and the secondhybrid coupler 20 provide the appropriate isolation between transmissionand receive.

FIG. 2 illustrates exemplary procedures that may be implemented toprovide RF duplexing. As explained in further detail below, theprocedures described in FIG. 2 are implemented by the tunable RFduplexer 10 shown in FIG. 1. Different embodiments of these exemplaryprocedures may be implemented depending on a particular componentstructure of a tunable RF duplexer 10. Furthermore, the order in whichthe procedures are presented is not intended to imply a requiredsequence for the procedures. Rather, the procedures may be implementedin a different sequence and/or some or all of the procedures may beimplemented simultaneously.

As shown in FIG. 2, the tunable RF duplexer 10 receives the RFtransmission input signal 26 from an upstream transceiver chain(procedure 1000). For example, the RF transmission input signal 26 isreceived by the tunable RF duplexer 10 from the power amplifier 28. Thepower amplifier 28 transfers power into the RF transmission input signal26 so that the RF transmission output signal 16 has a sufficiently highspectral density for external propagation from the antenna 12.Additionally, the tunable RF duplexer 10 receives the RF receive inputsignal 14 (procedure 1002). The RF receive input signal 14 was initiallyintercepted by the antenna 12.

FIG. 2A illustrates a graph of one embodiment of the RF receive inputsignal 14 and the RF transmission input signal 26 in the frequencydomain. Both the RF receive input signal 14 and the RF transmissioninput signal 26 operate in an RF communication band 34. In thisembodiment, the RF communication band 34 is the set of frequenciesbetween a cutoff frequency C1 and a cutoff frequency C2. Additionally,an RF receive band 36 and an RF transmission band 38 are defined withinthe RF communication band 34. The RF transmission band 38 is defined asthe set of frequencies between a cutoff frequency f_(TB1) and a cutofffrequency f_(TB2). Similarly, the RF receive band 36 is defined as theset of frequencies between a cutoff frequency f_(RB1) and a cutofffrequency f_(RB2). The RF receive input signal 14 operates in the RFreceive band 36 of the RF communication band 34.

In this example, a signal bandwidth 40 of the RF receive input signal 14is the set of frequencies that correspond to the portion of the RFreceive input signal 14 within 3 dB of a maximum magnitude 42. The RFreceive input signal 14 shown in FIG. 2A operates at a frequency f_(R).This frequency f_(R) corresponds to the maximum magnitude 42 of the RFreceive input signal 14. For example, the frequency f_(R) may be acarrier frequency of the RF receive input signal 14. The signalbandwidth 40 reaches a cutoff frequency f_(RBW1), since the frequencyf_(RBW1) corresponds to a value of the RF receive input signal 14 thatis 3 dB from the maximum magnitude 42. The signal bandwidth 40 alsoreaches a cutoff frequency f because the cutoff frequency f_(RBW2)corresponds to a value of the RF receive input signal 14 that is 3 dBfrom the maximum magnitude 42.

With regard to the RF transmission input signal 26, the RF transmissioninput signal 26 operates in the RF transmission band 38. Morespecifically, a signal bandwidth 43 of the RF transmission input signal26 is within the RF transmission band 38. A maximum magnitude 44 of theRF transmission input signal 26 is placed at the frequency f_(T). Acutoff frequency f_(TBW1) of the signal bandwidth 43 corresponds to avalue of the RF transmission input signal 26 at 3 dB from the maximummagnitude 44. Similarly, a cutoff frequency f_(TBW2) corresponds to avalue of the RF transmission input signal 26 that is 3 dB from themaximum magnitude 44. The RF transmission input signal 26 may be said tooperate at the frequency f_(T) since the frequency f_(T) is a carrier ora center frequency.

Portions of the RF receive input signal 14 and the RF transmission inputsignal 26 outside of their respective signal bandwidths 40, 43 may beconsidered spurious emissions. In other words, the portions of the RFreceive input signal 14 and the RF transmission input signal 26 may bereduced or eliminated without affecting the corresponding information ordata in the RF transmission input signal 26 and the RF receive inputsignal 14. Spurious emissions include parasitic emissions,intermodulation, interference, harmonic emissions, and frequencyconversion products. The signal bandwidth 40 and the signal bandwidth 43are defined as 3 dB bandwidths for pragmatic purposes. Generallyspeaking, at least for the types of signals being shown in FIG. 2A, thesignal bandwidths 40, 43 are measured by finding 3 dB magnitudes fromthe maximum magnitudes 42, 44, as explained above. However, moreaccurately, a necessary signal bandwidth is an exact amount of signalbandwidth required to carry the information or data of a signal.Anything outside of this necessary bandwidth would be consideredspurious emissions. Thus, the signal bandwidth 40 and the signalbandwidth 43 may or may not include a small portion of the spuriousemissions. The necessary signal bandwidths may be slightly smaller orslightly greater than the signal bandwidths 40 and 43.

Finally, it should be noted that the RF receive input signal 14 and theRF transmission input signal 26 shown in FIG. 2A are each narrow-bandsignals. Accordingly, the RF receive input signal 14 and the RFtransmission input signal 26 may represent time division multiplexing(TDM) signals, frequency division multiplexing (FDM) signals, spacedivision multiplexing (SDM) signals, and/or the like. Accordingly, thesenarrow-band signals may be said to operate at a particular frequency,which for the RF receive input signal 14 is the frequency f_(R) and forthe RF transmission input signal 26 is the frequency f_(T). The RFreceive band 36 is thus an RF receive channel within the RFcommunication band 34, while the RF transmission band 38 is an RFtransmission channel within the RF communication band 34.

However, this disclosure is not limited to narrow-band signals and theexamples given in FIG. 2A and throughout this disclosure are notintended to be limited in this manner. Rather, embodiments of thetunable RF duplexer 10 and the method shown in FIG. 2 may be providedfor wide-band signals, and also for both wide-band and narrow-bandsignals. With wide-band signals, such as orthogonal frequency divisionmultiple access (OFDMA) signals or code division multiple access (CDMA)signals, information or data is coded and spread across a larger portionof the spectrum. Thus, there would be no signal with a single carrierfrequency that has all of the information or data, but rather there maybe various carriers carrying different coded portions of theinformation. As such, the RF transmission band 38 for this type of RFtransmission input signal 26 may include various RF transmissionchannels. Similarly, the RF receive band 36 may include various RFreceive channels. With CDMA signals and other wide-band spectrumsignals, it is more practical to define the bandwidths by simply usingthe necessary bandwidth, as is known in the art.

FIG. 2B illustrates one embodiment of a frequency response 48 providedby the RF filter circuit 22. In FIG. 2B, the frequency response 48represents S21 and S12 parameters of each of the RF filters 23A, 23B(shown in FIG. 1) individually, as a function of frequency. Thus, thetwo-port S21 and S12 parameter of the RF filter 23A is represented bythe frequency response 48 in FIG. 2B. Since the RF filter 23B isidentical to the RF filter 23A, the two-port S21 and S12 parameter ofthe RF filter 23B is also represented by the frequency response 48 inFIG. 2B. The two-port S21 and S12 parameter represents the forward andreverse transmission of the RF filters 23A, 23B, as a function offrequency. A passband 50 corresponds to S21/S12 values in the frequencyresponse 48 that are within 3 dB of a maxima 51. A stopband 52 isdetermined relative to a minima 54. The maxima 51 and the minima 54 areset by the poles and zeros of the frequency response 48. The stopband 52is a set of frequencies that correspond to S21/S12 values within 3 dB ofthe minima 54. In this embodiment, the frequency response 48 defines thestopband 52 as a notch. As explained in further detail below, the RFfilter circuit 22 is tunable so as to shift the passband 50 and thestopband 52. Thus, by tuning the RF filter circuit 22, the frequencyresponse 48 may be transposed, so that the passband 50 and the stopband52 are provided at the desired frequency bands.

FIG. 2C illustrates the frequency response 48 once the passband 50 isshifted to include the RF receive band 36, and once the stopband 52 isshifted to include the RF transmission band 38. As shown in FIG. 2C, thetuning circuit 24 tunes the passband 50 so that the passband 50 includesthe RF receive band 36 (procedure 1004). The tuning circuit 24 may alsotune the stopband 52 so that the stopband 52 includes the RFtransmission band 38 (procedure 1006). In this manner, signals thatoperate in the RF receive band 36 are passed by the RF filter circuit22, while signals that operate in the RF transmission band 38 areblocked by the RF filter circuit 22.

FIG. 2D illustrates the frequency response 48 once the passband 50 isshifted to include the RF receive band 36, and once the stopband 52 isshifted to include the RF transmission band 38. However, in FIG. 2D, thefrequency response 48 represents S11 and S22 parameters of each of theRF filters 23A, 23B (shown in FIG. 1) individually, as a function offrequency. Thus, the two-port S11 and S22 parameter of the RF filter 23Ais represented by the frequency response 48 in FIG. 2D. Since the RFfilter 23B is identical to the RF filter 23A, the two-port S11 and S22parameter of the RF filter 23B is also represented by the frequencyresponse 48 in FIG. 2D. The two-port S11 and S22 parameter representsthe forward and reverse return loss of the RF filters 23A, 23B, as afunction of frequency. Note that in this embodiment, the S11/S22 valuesof the stopband 52 are at or near zero (0) dB in the RF transmissionband 38. By placing the stopband 52 at or near zero (0) dB, reflectionsin the stopband 52 are maximized, while insertion losses within the RFtransmission band 38 are minimized. Filtering thus removes noise outsideof the RF transmission band 38 while minimizing losses of reflectedsignals.

Referring now to FIGS. 2E and 2F, the RF receive input signal 14 issplit into a first RF quadrature hybrid receive signal (QHRS) 56 and asecond RF QHRS 58 (procedure 1008). As such, the first RF QHRS 56 is 90degrees or π/2 radians out of phase with the second RF QHRS 58. Also,the RF transmission input signal 26 is also split into a first RFquadrature hybrid transmission signal (QHTS) 60 and a second RF QHTS 62(procedure 1010). FIGS. 2E and 2F illustrate the first RF QHRS 56 andthe second RF QHRS 58. The first RF QHRS 56 and the second RF QHRS 58have substantially identical magnitude characteristics as the RF receiveinput signal 14. However, both the first RF QHRS 56 and the second RFQHRS 58 have a power spectral density that is at a power ratio withrespect to the power spectral density of the RF receive input signal 14.In this example, the power ratio is 3 dB, and thus the first RF QHRS 56and the second RF QHRS 58 have approximately one half of the power ofthe RF receive input signal 14. The first RF QHRS 56 and the second RFQHRS 58 are quadrature hybrids, since there is approximately a 90-degreeor π/2 radians phase difference between the signals. As explained infurther detail below, the first hybrid coupler 18 (FIG. 1) splits the RFreceive input signal 14 into the first RF QHRS 56 and the second RF QHRS58.

Of course, non-ideal characteristics of the tunable RF duplexer 10, suchas parasitic impedances, may result in the first RF QHRS 56 and thesecond RF QHRS 58 to be slightly unbalanced with respect to one another,or to have slightly less than half the power of the RF receive inputsignal 14. Also, non-ideal characteristics can result in the phasedifference between the first RF QHRS 56 and the second RF QHRS 58fluctuating somewhat from a 90-degree or π/2 radians phase difference.These types of errors are acceptable so long as the first RF QHRS 56 andthe second RF QHRS 58 can be combined into the RF receive output signal30, such that the RF receive output signal 30 complies with spectrumrequirements for the RF communication band 34.

Again referring to FIG. 2 and FIGS. 2E and 2F, the first hybrid coupler18 (FIG. 1) splits the RF transmission input signal 26 into the first RFQHTS 60 and the second RF QHTS 62. Both the first RF QHTS 60 and thesecond RF QHTS 62 have substantially identical magnitude characteristicsas the RF transmission input signal 26. However, a spectral density ofthe first RF QHTS 60 and the second RF QHTS 62 is at a power ratio withrespect to a power spectral density of the RF transmission input signal26. In this example, the power ratio is a 3 dB power ratio, and thus thefirst RF QHTS 60 and the second RF QHTS 62 are at about half power withrespect to the RF transmission input signal 26. The first RF QHTS 60 andthe second RF QHTS 62 are quadrature hybrids, because each has a90-degree or π/2 radians phase difference with respect to the other.Non-ideal characteristics of the first hybrid coupler 18, such asparasitic impedances, may result in the first RF QHTS 60 and the secondRF QHTS 62 having slightly lower power ratios with respect to the RFtransmission input signal 26. These non-ideal characteristics may alsoresult in the first RF QHTS 60 and the second RF QHTS 62 being slightlyunbalanced. Furthermore, the phase difference between the first RF QHTS60 and the second RF QHTS 62 may vary slightly due to non-idealcharacteristics of the tunable RF duplexer 10. These variations in bothpower and magnitude are acceptable, so long as the first RF QHTS 60 andthe second RF QHTS 62 can combine into the RF transmission output signal16 and comply with spectral requirements for the RF communication band34.

Since the RF receive input signal 14 is split into the first RF QHRS 56and the second RF QHRS 58, both the first RF QHRS 56 and the second RFQHRS 58 operate in the RF receive band 36 of the RF communication band34. As shown in FIGS. 2E and 2F, the frequency response 48 of the RFfilter circuit 22 has been transposed so that the passband 50 and thestopband 52 are shifted into the RF communication band 34. Moreparticularly, the passband 50 is shifted so that the RF receive band 36is in the passband 50. When the first hybrid coupler 18 outputs thefirst RF QHRS 56 and the second RF QHRS 58, the RF filter circuit 22filters the first RF QHRS 56 and the second RF QHRS 58 to pass the firstRF QHRS 56 and the second RF QHRS 58 within the passband 50 (procedure1012). Since the first RF QHRS 56 and the second RF QHRS 58 are withinthe passband 50, the RF filter circuit 22 is configured to pass thefirst RF QHRS 56 and the second RF QHRS 58 to the second hybrid coupler20.

However, the stopband 52 has been shifted so that the RF transmissionband 38 is in the stopband 52. When the second hybrid coupler 20 outputsthe first RF QHTS 60 and the second RF QHTS 62, the RF filter circuit 22filters the first RF QHTS 60 and the second RF QHTS 62 to reflect thefirst RF QHTS 60 and the second RF QHTS 62 within the stopband 52(procedure 1014). The first RF QHTS 60 and the second RF QHTS 62 operatewithin the RF transmission band 38. Since the stopband 52 is in the RFtransmission band 38, the RF filter circuit 22 has been tuned to blockthe first RF QHTS 60 and the second RF QHTS 62.

As shown in FIGS. 2E and 2F, the first RF QHRS 56 has a signal bandwidth64 and the second RF QHRS 58 has a signal bandwidth 66. Although thefirst RF QHRS 56 and the second RF QHRS 58 have approximately half thepower spectral density (excluding losses) of the RF receive input signal14 shown in FIG. 2A, the signal bandwidth 64 of the first RF QHRS 56 andthe signal bandwidth 66 of the second RF QHRS 58 are approximately thesame size and are located at approximately the same frequencies. InFIGS. 2E and 2F, the signal bandwidth 64 of the first RF QHRS 56 and thesignal bandwidth 66 of the second RF QHRS 58 are between the frequenciesf_(RBW1) and f_(RBW2). Both the signal bandwidth 64 and the signalbandwidth 66 are approximately equal to the signal bandwidth 40 (FIG.2A) of the RF receive input signal 14. However, non-ideal circuitbehavior may result in either slight misalignments and/or slight sizedifferences in the signal bandwidth 64 and the signal bandwidth 66. Theamount of error that is permissible may depend on the particularapplication and spectrum requirements. As mentioned above, the tuningcircuit 24 is configured to tune the frequency response 48 of the RFfilter circuit 22 so that the signal bandwidth 64 of the first RF QHRS56 and the signal bandwidth 66 of the second RF QHRS 58 are each withinthe passband 50. Consequently, the RF filter circuit 22 filters thefirst RF QHRS 56 and the second RF QHRS 58 to pass the first RF QHRS 56and the second RF QHRS 58 within the passband 50 to the second hybridcoupler 20.

As shown in FIGS. 2E and 2F, the first RF QHTS 60 has a signal bandwidth68, while the second RF QHTS 62 has a signal bandwidth 70. Although thefirst RF QHTS 60 and the second RF QHTS 62 have approximately half thepower spectral density of the RF transmission input signal 26 (excludinglosses), the signal bandwidth 68 of the first RF QHTS 60 and the signalbandwidth 70 of the second RF QHTS 62 are configured to be approximatelythe same as the signal bandwidth 43 of the RF transmission input signal26. In the embodiments illustrated in FIGS. 2E and 2F, the signalbandwidth 68 and the signal bandwidth 70 are both between thefrequencies f_(TBW1) and f_(TBW2). However, non-ideal circuit behaviormay result in slight misalignments or size differences in the signalbandwidth 68 and the signal bandwidth 70. The amount of error that ispermissible may depend on the particular application and spectrumrequirements. The tuning circuit 24 is configured to tune the frequencyresponse 48 of the RF filter circuit 22 so that the signal bandwidth 68of the first RF QHTS 60 and the signal bandwidth 70 of the second RFQHTS 62 are each within the stopband 52. In this manner, the RF filtercircuit 22 blocks the first RF QHTS 60 and the second RF QHTS 62 suchthat the first RF QHTS 60 and the second RF QHTS 62 are reflected backto the first hybrid coupler 18.

Referring now to FIGS. 2 and 2G, FIG. 2G illustrates one embodiment ofthe RF receive output signal 30 and one embodiment of the RFtransmission output signal 16. To provide the RF receive output signal30, the second hybrid coupler 20 combines the first RF QHRS 56 (shown inFIG. 2E) and the second RF QHRS 58 (shown in FIG. 2F) within thepassband 50 (shown in FIGS. 2E and 2F) into the RF receive output signal30 (procedure 1016). As shown in FIG. 2G, the RF receive output signal30 operates within the RF receive band 36. Ideally, the RF receiveoutput signal 30 has the same power spectral density as the RF receiveinput signal 14. However, in practice, the tunable RF duplexer 10 maynot be lossless, and thus the power spectral density of the RF receiveoutput signal 30 may be somewhat smaller than the power spectral densityof the RF receive input signal 14.

As shown in FIG. 2G, the RF receive output signal 30 has a signalbandwidth 72 within the RF receive band 36. In this example, the signalbandwidth 72 is the same as the signal bandwidth 40 (shown in FIG. 2A)of the RF receive input signal 14. Thus, the signal bandwidth 72 isbetween the frequencies f_(RBW1) and f_(RBW2). Furthermore, the RFreceive output signal 30 operates at the RF receive frequency f_(R),like the RF receive input signal 14. However, this may not be exactlythe case in all circumstances. For example, the signal bandwidth 72 maybe slightly smaller, larger, or misaligned with the signal bandwidth 40of the RF receive input signal 14. There may also be a small frequencydisplacement between the RF receive frequency of the RF receive inputsignal 14 and the RF receive frequency f_(R) of the RF receive outputsignal 30. The amount of error permissible for these parameters maydepend on the particular application and/or spectrum requirements.Furthermore, additional circuitry may be provided in order to compensatefor these errors.

Referring again to FIGS. 2 and 2G, FIG. 2G also illustrates oneembodiment of the RF transmission output signal 16. To provide the RFtransmission output signal 16, the first hybrid coupler 18 combines thefirst RF QHTS 60 (shown in FIG. 2E) and the second RF QHTS 62 (shown inFIG. 2F) within the stopband 52 (shown in FIGS. 2E and 2F) into the RFtransmission output signal 16 (procedure 1018). Ideally, the RFtransmission output signal 16 has the same power spectral density as theRF transmission input signal 26 shown in FIG. 2A. However, the tunableRF duplexer 10 may not be lossless, and thus, the power spectral densityof the RF transmission output signal 16 may be smaller. As shown in FIG.2G, the RF transmission output signal 16 has a signal bandwidth 74. Thesignal bandwidth 74 may be the same as the signal bandwidth 43 (shown inFIG. 2A) of the RF transmission input signal 26. Thus, in FIG. 2G, thesignal bandwidth 74 of the RF transmission output signal 16 is betweenthe frequency f_(TBW1) and f_(TBW2). However, this may not be the case.For example, the signal bandwidth 74 may be smaller, larger, ormisaligned with the signal bandwidth 43. Also, the RF transmissionoutput signal 16 shown in FIG. 2G operates at the RF transmissionfrequency f_(T), like the RF transmission input signal 26. This may ormay not be the case. For example, there may be a small displacementbetween the RF transmission frequency f_(T) of the RF transmissionoutput signal 16 and the RF transmission frequency f_(T) of the RFtransmission input signal 26. The amount of error that is permissible inthese parameters may depend on a particular application and/or spectrumrequirements.

FIG. 3A illustrates a tunable RF duplexer 10A along with a receivesignal flow of the tunable RF duplexer 10A. The tunable RF duplexer 10Aincludes the first hybrid coupler 18, the second hybrid coupler 20, anRF filter circuit 22A, and a tuning circuit 24A. As shown in FIG. 3A,the first hybrid coupler 18 has a first port 76, a second port 78, athird port 80, and a fourth port 82, while the second hybrid coupler 20has a fifth port 84, a sixth port 86, a seventh port 88, and an eighthport 90.

The antenna 12 intercepts the RF receive input signal 14 aselectromagnetic waves in free space. These electromagnetic waves resultin excitations within the antenna 12, thereby converting theelectromagnetic waves into the RF receive input signal 14. The firsthybrid coupler 18 is operable to receive the RF receive input signal 14.In this particular embodiment, the first hybrid coupler 18 is coupled toreceive the RF receive input signal 14 at the first port 76 from theantenna 12. The first hybrid coupler 18 is operable to split the RFreceive input signal 14 into the first RF QHRS 56 and the second RF QHRS58. In this manner, the first RF QHRS 56 and the second RF QHRS 58 haveapproximately the same power ratio with respect to the RF receive inputsignal 14, but have a quadrature phase difference of approximately 90degrees or π/2 radians with respect to one another.

With regard to the first hybrid coupler 18 shown in FIG. 3A, the firstport 76 is phase-aligned with the third port 80, while the first port 76has a quadrature phase shift with respect to the fourth port 82. Thus,the first RF QHRS 56 at the third port 80 is approximately phase-alignedwith the RF receive input signal 14 at the first port 76, but there is aquadrature phase difference between the RF receive input signal 14 atthe first port 76 and the second RF QHRS 58 at the fourth port 82.

Note that in alternative embodiments, this may or may not be the case.For example, there may be a phase shift between the first port 76 andthe third port 80 (such as +45 degrees or +π/4 radians). The phase shiftbetween the first port 76 and the fourth port 82 may then be equal tothis phase shift plus or minus 90 degrees (such as +135 degrees or +3π/4radians, −45 degrees or −π/4 radians). Accordingly, so long as the phasedifference between the first RF QHRS 56 and the second RF QHRS 58 isabout 90 degrees or π/2 radians, phase alignment between the third port80 and the first port 76, and between the fourth port 82 and the firstport 76, can vary.

The first RF QHRS 56 is output at the third port 80 to the RF filtercircuit 22A. Additionally, the second RF QHRS 58 is output at the fourthport 82 to the RF filter circuit 22A. In this embodiment, the RF filtercircuit 22A has a first RF filter 92 (an embodiment of the RF filter 23Ashown in FIG. 1) and a second RF filter 94 (an embodiment of the RFfilter 23B shown in FIG. 1). The first RF filter 92 is coupled to thethird port 80 so as to receive the first RF QHRS 56 from the firsthybrid coupler 18. The second RF filter 94 is coupled to the fourth port82 so as to receive the second RF QHRS 58 from the first hybrid coupler18. The first RF filter 92 and the second RF filter 94 each have afrequency response. The frequency response of the RF filter circuit 22Ais thus determined in accordance with the combined effect of theindependent frequency responses provided by the first RF filter 92 andthe second RF filter 94. However, in this embodiment, the first RFfilter 92 and the second RF filter 94 are identical to one another. Assuch, the overall frequency response of the RF filter circuit 22A is thesame as the independent frequency responses provided by the first RFfilter 92 or the second RF filter 94. Alternatively, in otherembodiments, the first RF filter 92 and the second RF filter 94 may bedifferent and/or may be tuned independently by the tuning circuit 24A.As such, the different independent frequency responses from the first RFfilter 92 and the second RF filter 94 may combine to determine theoverall frequency response of the RF filter circuit 22A.

Referring again to FIG. 3A, the tuning circuit 24A is configured to tunethe frequency response of the RF filter circuit 22A so that the passbandincludes the RF receive band. The tuning circuit 24A thus shifts thepassband of the RF filter circuit 22A to include the RF receive band. Inthis manner, the RF filter circuit 22A is operable to pass the first RFQHRS 56 and the second RF QHRS 58 to the second hybrid coupler 20. Themanner of tuning the frequency response may depend on the topology ofthe RF filter circuit 22A. For example, the first RF filter 92 and thesecond RF filter 94 shown in FIG. 3A may both be passive filters.Accordingly, one or more reactive impedance components (inductive,capacitive, or both) in each of the first and second RF filters 92, 94may have a variable reactive impedance level. By varying these variablereactive impedance levels, the poles and zeros of the individualfrequency responses provided by each of the first and second RF filters92, 94 are adjusted. This thereby shifts the passband and/or thestopband of the RF filter circuit 22A. Ideally, the first RF filter 92and the second RF filter 94 in FIG. 3A each operate as a short withrespect to the RF receive band and as an open circuit with respect tothe RF transmission band.

The tuning circuit 24A illustrated in FIG. 3A generates a receive tuningcontrol output 96 and a transmission tuning control output 98. Thevariable reactive impedance components in both the first RF filter 92and the second RF filter 94 are set in accordance with one or moresignal levels of the receive tuning control output 96. In this manner,the stopband is shifted to include the RF transmission band inaccordance with one or more signal levels of the transmission tuningcontrol output 98. Similarly, reactive impedance levels of variablereactive components in the first RF filter 92 and the second RF filter94 are set in accordance with one or more signal levels of the receivetuning control output 96. The RF filter circuit 22A may also includeactive RF filters, Surface Acoustic Wave (SAW) filters, or any othertype of RF filter or combination of RF filters that is suitable toprovide a desired frequency response. As such, the tuning circuit 24Amay employ various types of tuning topologies, depending on theparticular filtering topology being employed by the RF filter circuit22A.

By placing the passband in the RF receive band, the RF filter circuit22A passes the first RF QHRS 56 and the second RF QHRS 58 to the secondhybrid coupler 20. In this particular embodiment, the first RF filter 92passes the first RF QHRS 56 to the second hybrid coupler 20, while thesecond RF filter 94 passes the second RF QHRS 58 to the second hybridcoupler 20. As mentioned above, the passband is set in the RF receiveband.

Referring again to FIG. 3A, the second hybrid coupler 20 receives thefirst RF QHRS 56 from the first RF filter 92 at the fifth port 84. Thesecond RF QHRS 58 is received by the second hybrid coupler 20 from thesecond RF filter 94 at the sixth port 86. As discussed above, the firstRF QHRS 56 and the second RF QHRS 58 have a quadrature phase differenceof about 90 degrees or π/2 radians. Thus, for example, if the first RFQHRS 56 has a phase of zero degrees, the second RF QHRS 58 would have aphase of approximately 90 degrees (or π/2 radians). From the fifth port84 to the seventh port 88, the second hybrid coupler 20 provides nophase shift. Alternatively, the second hybrid coupler 20 may beconfigured to provide a phase shift from the fifth port 84 to theseventh port 88 of Δ (i.e., such as +45 degrees or π/4 radians).

The second hybrid coupler 20 shown in FIG. 3A is configured to provide aquadrature phase shift from the sixth port 86 to the seventh port 88. Inthis example, the phase shift is 90 degrees (or π/2 radians), and thusthe second RF QHRS 58 has a phase, as seen from the seventh port 88, of180 degrees (note that the second RF QHRS 58 was received with a phaseof 90 degrees in this example, and thus is seen with a phase of 180degrees with another phase shift of 90 degrees). Alternatively, thephase shift from the sixth port 86 to the seventh port 88 may be Δ±90degrees (or π/2 radians). In any case, the phase difference between thefirst RF QHRS 56 and the second RF QHRS 58 as seen from the seventh port88 is about 180 degrees (note that the first RF QHRS 56 was receivedwith a phase of zero (0) degrees and thus is seen with a phase of zero(0) degrees at the seventh port 88). Accordingly, the quadrature phaseshift at the seventh port 88 from the sixth port 86 results indestructive interference between the first RF QHRS 56 and the second RFQHRS 58 at the seventh port 88. As a result, the first RF QHRS 56 andthe second RF QHRS 58 are substantially cancelled at the seventh port88. In this manner, the seventh port 88 is substantially isolated fromreceive signal flow.

The second hybrid coupler 20 is configured to output the RF receiveoutput signal 30 from the eighth port 90 in response to the first RFQHRS 56 being received from the RF filter circuit 22A at the fifth port84 and the second RF QHRS 58 being received from the RF filter circuit22A at the sixth port 86. In this particular embodiment, the secondhybrid coupler 20 is configured to pass the second RF QHRS 58 from thesixth port 86 to the eighth port 90. The second hybrid coupler 20provides no phase shift to the second RF QHRS 58 from the sixth port 86to the eighth port 90. The second RF QHRS 58 is thus passed with a phaseof 90 degrees to the eighth port 90. Alternatively, the second hybridcoupler 20 may provide a phase shift of Δ (i.e., such as +45 degrees orπ/4 radians) to the second RF QHRS 58 when passed from the sixth port 86to the eighth port 90. The second hybrid coupler 20 is configured topass the first RF QHRS 56 from the fifth port 84 to the eighth port 90.The second hybrid coupler 20 provides a quadrature phase shift to thefirst RF QHRS 56 at the eighth port 90. In this embodiment, thequadrature phase shift is 90 degrees or π/2 radians. Alternatively, if aphase shift of Δ was provided to the second RF QHRS 58 from the sixthport 86 to the eighth port 90, the quadrature phase shift would be Δ±90degrees (or π/2 radians).

Accordingly, the first RF QHRS 56 is provided substantially as aduplicate of the second RF QHRS 58 at the eighth port 90. This is aresult of the quadrature phase shift provided to the first RF QHRS 56when passed from the fifth port 84 to the eighth port 90 (now, at theeighth port 90, the first RF QHRS 56 is shifted to have a phase of 90degrees, just like the second RF QHRS 58). Referring again to theexample discussed previously, if the first RF QHRS 56 has a phase ofzero at the fifth port 84, then the first RF QHRS 56 has a phase of 90degrees at the eighth port 90. If the phase of the second RF QHRS 58 atthe sixth port 86 is 90 degrees, then the phase of the second RF QHRS 58is also 90 degrees at the eighth port 90. Accordingly, the first RF QHRS56 is substantially a duplicate of the second RF QHRS 58 at the eighthport 90 because the first RF QHRS 56 and the second RF QHRS 58 becomephase-aligned at the eighth port 90. As a result, the first RF QHRS 56and the second RF QHRS 58 constructively interfere at the eighth port 90to output the RF receive output signal 30 from the eighth port 90. Notethat since the first RF QHRS 56 and the second RF QHRS 58 substantiallycancel at the seventh port 88 due to destructive interference, verylittle or no power is transferred from the first RF QHRS 56 and thesecond RF QHRS 58 to the seventh port 88. Instead most, if not all, ofthe power in the first RF QHRS 56 and the second RF QHRS 58 istransferred to the eighth port 90 and provided in the RF receive outputsignal 30.

An impedance load 100 is coupled to the seventh port 88 of the secondhybrid coupler 20. The impedance load 100 may be a 50 Ohm load. Due tothe phase-shifting provided by the first hybrid coupler 18 and thesecond hybrid coupler 20, spurious emissions from the second port 78 ofthe first hybrid coupler 18 would see a very high (theoreticallyinfinite) impedance level at the eighth port 90 of the second hybridcoupler, but only the impedance load 100 at the seventh port 88. Thus,the spurious emissions are aggregated to be an aggregated noise signal102 at the seventh port 88. This aggregated noise signal 102 isdissipated by the impedance load 100. Additionally, the eighth port 90is isolated from the seventh port 88. As such, the seventh port 88 issubstantially unresponsive to signals incident at the eighth port 90 andthe seventh port 88 is substantially unresponsive to signals incident atthe eighth port 90.

Additionally, an antenna impedance tuner 104 is coupled between theantenna 12 and the first port 76. Since the RF transmission outputsignal 16 is in the RF transmission band, the impedance of the antenna12 may result in a portion of the RF transmission output signal 16 beingreflected back to the first port 76. This not only would degrade the RFtransmission output signal 16, but would also degrade the isolationbetween the receive signal flow and the eighth port 90. The antennaimpedance tuner 104 is tunable so as to reduce reflections from theantenna 12 to the first port 76. More specifically, an impedance of theantenna 12 may be tuned so as to provide impedance matching between thefirst port 76 and the antenna 12.

FIG. 3B illustrates the tunable RF duplexer 10A along with thetransmission signal flow. The RF filter circuit 32A may be coupled tothe power amplifier 28 to receive the RF transmission input signal 26.The RF filter circuit 32A may be tunable so as to define a stopband. Thetuning circuit 24A may be configured to tune the RF filter circuit 32Ato provide the stopband within the RF receive band. In particular, theRF filter circuit 32A is operable to receive the transmission tuningcontrol output 98 from the tuning circuit 24A, which tunes the stopbandaccordingly. As a result, the RF filter circuit 32A is operable tofilter out spurious emissions resulting from upstream RF circuitry, suchas the power amplifier 28, that are within the RF receive band.

The first hybrid coupler 18 is operable to receive the RF transmissioninput signal 26 at the second port 78. The RF transmission input signal26 operates in the RF transmission band, which is within the same RFcommunication band as the RF receive band of the RF receive input signal14 (see FIG. 2A). The first hybrid coupler 18 is operable to split theRF transmission input signal 26 into the first RF QHTS 60 and the secondRF QHTS 62. Since the first RF QHTS 60 and the second RF QHTS 62 arequadrature hybrids, the first RF QHTS 60 and the second RF QHTS 62 areapproximately equal in power, but have a quadrature phase difference of90 degrees or π/2 radians. The first hybrid coupler 18 is operable tooutput the first RF QHTS 60 from the fourth port 82 and to output thesecond RF QHTS 62 from the third port 80 in response to receiving the RFtransmission input signal 26 at the second port 78.

In the embodiment illustrated in FIG. 3B, the first RF QHTS 60 isphase-aligned with the RF transmission input signal 26, while the secondRF QHTS 62 has a phase difference of about 90 degrees with respect tothe RF transmission input signal 26. It should be noted that this may ormay not be the case. For example, in alternative embodiments, a phaseshift of Δ (i.e., +45 degrees or π/4 radians) may be provided betweenthe second port 78 and the fourth port 82, and thus, a phase shift ofΔ±90 degrees (or π/2 radians) would be provided between the second port78 and the fourth port 82.

The RF filter circuit 22A is operable to reflect the first RF QHTS 60and the second RF QHTS 62. As discussed above, the frequency response ofthe RF filter circuit 22A defines the stopband and the RF filter circuit22A is tunable so as to shift the stopband. For example, the stopbandmay be a notch that is shiftable. The tuning circuit 24A is configuredto tune the frequency response of the RF filter circuit 22A so that thesignal bandwidth of the first RF QHTS 60 and the signal bandwidth of thesecond RF QHTS 62 are each within the stopband. For instance, the tuningcircuit 24A may be configured to place the notch within the RFtransmission band so that the notch is centered at the RF transmissionsignal frequency. In this embodiment, the tuning circuit 24A generatesthe transmission tuning control output 98. Variable reactive impedancecomponents in both the first RF filter 92 and the second RF filter 94are responsive to the one or more signal levels of the transmissiontuning control output 98 so as to adjust the variable impedance levelsbased on the signal level(s) of the transmission tuning control output98. As a result, the notch defined by the individual frequency responseof the first RF filter 92 is shifted to include the signal bandwidth ofthe second RF QHTS 62. Also, the notch defined by the individualfrequency response of the second RF filter 94 is shifted to include thesignal bandwidth of the first RF QHTS 60. In other words, the notchesdefined by the individual frequency responses of the first RF filter 92and the second RF filter 94 are placed in the RF transmission band.

Since the tuning circuit 24A has tuned the frequency response of the RFfilter circuit 22A so that the stopband includes the RF transmissionsignal band, the RF filter circuit 22A blocks the first RF QHTS 60 andthe second RF QHTS 62. Accordingly, the second hybrid coupler 20 issubstantially isolated from the transmission signal flow. The RF filtercircuit 22A then reflects the first RF QHTS 60 and the second RF QHTS 62back to the first hybrid coupler 18. In the embodiment illustrated inFIG. 3B, the second RF filter 94 reflects the first RF QHTS 60 back tothe first hybrid coupler 18 at the fourth port 82. The first RF filter92 reflects the second RF QHTS 62 back to the first hybrid coupler 18 atthe third port 80.

The first hybrid coupler 18 is configured to combine the first RF QHTS60 and the second RF QHTS 62 into the RF transmission output signal 16.To combine the first RF QHTS 60 and the second RF QHTS 62 into the RFtransmission output signal 16, the first hybrid coupler 18 is configuredto pass the second RF QHTS 62 from the fourth port 82 to the first port76. Additionally, the first hybrid coupler 18 is configured to pass thefirst RF QHTS 60 from the fourth port 82 to the first port 76. However,the first hybrid coupler 18 provides a quadrature phase shift to thefirst RF QHTS 60 from the fourth port 82 to the first port 76. Thus, thefirst RF QHTS 60 is provided substantially as a duplicate of the secondRF QHTS 62 at the first port 76. For example, if the phase of the secondRF QHTS 62 is 90 degrees at the third port 80, the second RF QHTS 62 hasa phase of 90 degrees at the first port 76. Additionally, the phase ofthe first RF QHTS 60 at the third port 80 is approximately zero degrees.However, due to the quadrature phase shift between the fourth port 82and the first port 76, the first RF QHTS 60 has a phase of about 90degrees at the first port 76. Accordingly, the first RF QHTS 60 and thesecond RF QHTS 62 constructively interfere at the first port 76 tooutput the RF transmission output signal 16 from the first port 76.

Also, note that the first hybrid coupler 18 is configured such that thequadrature phase shift at the second port 78 results in destructiveinterference between the first RF QHTS 60 and the second RF QHTS 62.Referring again to the previous example provided, at the second port 78,the first RF QHTS 60 appears to have a phase of zero degrees, but thesecond RF QHTS 62 appears to have a phase of 180 degrees. As a result,the first RF QHTS 60 and the second RF QHTS 62 are substantiallycancelled at the second port 78. Consequently, most, if not all, of thepower of the first RF QHTS 60 and the second RF QHTS 62 is transferredto the first port 76 and provided in the RF transmission output signal16. The first hybrid coupler 18 is thus configured to output the RFtransmission output signal 16 from the first port 76 in response to thefirst RF QHTS 60 being reflected back by the RF filter circuit 22A tothe fourth port 82 and the second RF QHTS 62 being reflected back by theRF filter circuit 22A to the third port 80.

Note that spurious emissions from the first RF QHTS 60 and the second RFQHTS 62 that are not reflected by the RF filter circuit 22A, such asspurious noise emissions outside the stopband, may be passed toward thesecond hybrid coupler 20. In this example, the spurious emissions woulddestructively interfere at the eighth port 90 and constructivelyinterfere at the seventh port 88. Accordingly, these spurious emissionsbecome part of the aggregated noise signal 102 and are dissipated by theimpedance load 100. Therefore, the tunable RF duplexer 10A significantlyreduces noise interference resulting at the second port 78 of the firsthybrid coupler 18 from degrading the RF receive output signal 30 at theeighth port 90.

As shown in FIGS. 3A and 3B, the tunable RF duplexer 10A allows thereceive signal flow and the transmission signal flow to be simultaneous.More specifically, the first port 76 of the first hybrid coupler 18 isconfigured to simultaneously receive the RF receive input signal 14 andoutput the RF transmission output signal 16 to and from the antenna 12.As a result, the RF transmission output signal 16 is output from thefirst port 76 while the RF receive input signal 14 is being received atthe first port 76. The tunable RF duplexer 10A can provide thisfunctionality because the second port 78 is substantially isolated fromthe eighth port 90.

Note that the first RF filter 92 in the RF filter circuit 22A is coupledin series between the third port 80 of the first hybrid coupler 18 andthe fifth port 84 of the second hybrid coupler 20. Furthermore, thesecond RF filter 94 in the RF filter circuit 22A is coupled in seriesbetween the fourth port 82 of the first hybrid coupler 18 and the sixthport 86 of the second hybrid coupler 20. As noted above, the stopband ofthe RF filter circuit 22A, and thus the stopband of each of the firstand second RF filters 92, 94, should reflect the first RF QHTS 60 andthe second RF QHTS 62. As a result, an impedance of the second RF filter94 should appear very high within the RF transmission band as seen fromthe fourth port 82, and an impedance of the first RF filter 92 shouldappear very high within the RF transmission band as seen from the thirdport 80. Ideally, the impedance of the second RF filter 94 as seen fromthe fourth port 82 and the impedance of the first RF filter 92 as seenfrom the third port 80 should be infinite. In practice, these impedancesshould simply be high enough to meet RF communication bandspecifications.

Additionally, the passband of the RF filter circuit 22A, and thus thepassband of each of the first and second RF filters 92, 94, should passthe first RF QHRS 56 and the second RF QHRS 58. The first and second RFfilters 92, 94 should therefore appear transparent in the RF receiveband. Since the first and second RF filters 92, 94 are coupled inseries, the input/output impedances of each of the first and second RFfilters 92, 94 would ideally be the same as the characteristic impedanceof the intra-hybrid connection lines. In practice, the input/outputimpedances of each of the first and second RF filters 92, 94 may simplybe provided sufficiently near the characteristic impedance of theintra-hybrid connection lines such that RF communication bandspecifications are met. Nevertheless, tuning by the tuning circuit 24Acan be difficult to implement when the first and second RF filters 92,94 are coupled in series, as shown in FIGS. 3A and 3B.

Referring now to FIG. 4, FIG. 4 illustrates one embodiment of afrequency response 48A provided by the RF filter circuit 22A shown inFIGS. 3A and 3B. The frequency response 48A defines a passband 50A and astopband 52A. The RF filter circuit 22A described in FIGS. 3A and 3B istunable so that the RF communication band (referred to generically aselement 34 and specifically as elements 34A-34E) can be switched to beany one of a plurality of different RF communication bands 34A-34E. Thedifferent RF communication bands 34A-34E may each define different RFreceive band(s) and different RF transmission band(s) within each of theRF communication bands 34A-34E. Accordingly, the RF transmission inputsignal 26 (shown in FIGS. 3A-3B) and the RF receive input signal 14(shown in FIGS. 3A-3B) may be in the respective RF receive band and RFtransmission band of any of the different RF communication bands34A-34E. Accordingly, the first RF QHRS 56, the second RF QHRS 58, andthe RF receive output signal 30 would also operate within the RF receiveband of the RF receive input signal 14. Similarly, the first RF QHTS 60,the second RF QHTS 62, and the RF transmission output signal 16 wouldoperate in the respective RF transmission band of the RF transmissioninput signal 26.

When the RF communication band 34 is switched to be a different one ofthe plurality of RF communication bands 34A-34E, the tuning circuit 24Ais operable to tune the frequency response 48A to the particular RFcommunication band 34A-34E. Since the RF communication band 34 has beenswitched, the first RF QHRS 56 and the second RF QHRS 58 are provided inthe new RF communication band 34A-34E. In particular, the first RF QHRS56 and the second RF QHRS 58 operate within the RF receive band of thenew RF communication band 34A-34E. Also, since the RF communication band34 has been switched to the new RF communication band 34A-34E, the firstRF QHTS 60 and the second RF QHTS 62 are provided within the new RFcommunication band 34A-34E. In particular, the first RF QHTS 60 and thesecond RF QHTS 62 operate within the RF transmission band of the new RFcommunication band 34A-34E. The tuning circuit 24A tunes the passband50A so as to pass the first RF QHRS 56 and the second RF QHRS 58 inresponse to the RF communication band 34 being switched to the new RFcommunication band 34A-34E. More particularly, the passband 50A isprovided so as to include the RF receive band of the selected RFcommunication band 34A-34E. Similarly, the tuning circuit 24A shifts thestopband 52A to reflect the first RF QHTS 60 and the second RF QHTS 62in response to the RF communication band 34 being switched to the new RFcommunication band 34A-34E. More particularly, the stopband 52A isshifted to include the RF transmission band of the selected RFcommunication band 34A-34E.

FIG. 5 illustrates another embodiment of a tunable RF duplexer 10B. Thetunable RF duplexer 10B is similar to the tunable RF duplexer 10A shownin FIGS. 3A and 3B. The tunable RF duplexer 10B has a first hybridcoupler 18′ and a second hybrid coupler 20′, an RF filter circuit 22B,and a tuning circuit 24B. The RF filter circuit 22B is anotherembodiment of the RF filter circuit 22 in FIG. 1. As explained infurther detail below, the first hybrid coupler 18′ and the second hybridcoupler 20′ operate in the same manner as the first hybrid coupler 18and second hybrid coupler 20 described above in FIGS. 3A and 3B, exceptin this embodiment the first hybrid coupler 18′ and the second hybridcoupler 20′ are tunable. Also, like the RF filter circuit 22A shown inFIGS. 3A and 3B, the RF filter circuit 22B has a frequency response thatdefines a stopband and a passband and is tunable so as to shift thepassband and the stopband. However, a first RF filter 106 has anindividual frequency response that defines a passband and a stopband.Unlike the RF filter circuit 22A shown in FIGS. 3A and 3B, the RF filtercircuit 22B in FIG. 5 has the first RF filter 106 and a second RF filter108 coupled in shunt. More specifically, the first RF filter 106 iscoupled in shunt between the third port 80 and the fifth port 84, whilethe second RF filter 108 is coupled in shunt between the fourth port 82and the sixth port 86.

The tuning circuit 24B illustrated in FIG. 5 is configured to tune theRF filter circuit 22B so as to shift the passband to within the RFreceive band of the RF receive input signal 14 and the stopband towithin the RF transmission input signal 26. More specifically, the firstRF filter 106 has an individual frequency response that defines apassband and a stopband. The second RF filter 108 also has an individualfrequency response that defines a passband and a stopband. As such, thedifferent independent frequency responses from the first RF filter 106and the second RF filter 108 may combine to determine the overallfrequency response of the RF filter circuit 22B. Alternatively, in otherembodiments, the first RF filter 106 and the second RF filter 108 may bedifferent, and/or may be tuned independently by the tuning circuit 24B.As such, the different independent frequency responses from the first RFfilter 106 and the second RF filter 108 may combine to determine theoverall frequency response of the RF filter circuit 22B.

The tuning circuit 24B shown in FIG. 5 is configured to generate areceive tuning control output 110, hybrid control outputs (referred togenerically as element 111 and specifically as elements 111A and 111B),and a transmission tuning control output 112. The variable reactiveimpedance components of both the first RF filter 106 and the second RFfilter 108 are set in accordance with one or more signal levels of thereceive tuning control output 110. In this manner, the passbands of boththe first RF filter 106 and the second RF filter 108 are shifted withinthe RF receive band of the RF receive input signal 14. The second hybridcoupler 20′ is operable to output the RF receive output signal 30 at theeighth port 90. Additionally, reactive impedance levels of variablereactive components in both the first RF filter 106 and the second RFfilter 108 are set in accordance with a signal level of the transmissiontuning control output 112. In this manner, the stopbands of both thefirst RF filter 106 and the second RF filter 108 are shifted within theRF transmission band of the RF transmission input signal 26 at thesecond port 78. The first hybrid coupler 18′ is configured to output theRF transmission output signal 16 at the first port 76. As such, thetunable RF duplexer 10B is configured to simultaneously receive the RFreceive input signal 14 from the antenna 12 and output the RFtransmission output signal 16 to the antenna 12. The tuning of the firsthybrid coupler 18′ by the first hybrid control output 111A and thetuning of the second hybrid coupler 20′ by the second hybrid controloutput 111B is described in further detail below.

FIG. 6A illustrates the tunable RF duplexer 10B along with a receivesignal flow. The antenna 12 intercepts the RF receive input signal 14 aselectromagnetic waves in free space. These electromagnetic waves resultin excitations within the antenna 12, thereby converting theelectromagnetic waves into the RF receive input signal 14. The firsthybrid coupler 18′ is operable to receive the RF receive input signal 14at the first port 76.

The first hybrid coupler 18′ is operable to split the RF receive inputsignal 14 into the first RF QHRS 56 and the second RF QHRS 58. In thismanner, the first RF QHRS 56 and the second RF QHRS 58 haveapproximately the same power ratio with respect to the RF receive inputsignal 14, but have a quadrature phase difference of approximately 90degrees or π/2 radians with respect to one another. The first RF QHRS 56is output at the third port 80 to the RF filter circuit 22B.Additionally, the second RF QHRS 58 is output at the fourth port 82 tothe RF filter circuit 22B.

The tuning circuit 24B is configured to tune the frequency response sothat a signal bandwidth of the first RF QHRS 56 and a signal bandwidthof the second RF QHRS 58 are each within the passband of the RF filtercircuit 22B. In this embodiment, the tuning circuit 24B is configured totune the passband of the first RF filter 106 within the signal bandwidthof the first RF QHRS 56 and to tune the passband of the second RF filter108 within the signal bandwidth of the second RF QHRS 58. To provideeach of the passbands, the first RF filter 106 and the second RF filter108 each appear approximately as open circuits. Since the first RFfilter 106 is coupled in shunt, the first RF filter 106 appearsapproximately transparent within the passband of the first RF filter106. Similarly, since the second RF filter 108 is coupled in shunt, thesecond RF filter 108 appears approximately transparent within thepassband of second RF filter 108. Unlike the series coupled first andsecond RF filters 92, 94 shown in FIGS. 3A and 3B, the input/outputimpedances of the first RF filter 106 and the second RF filter 108 inFIG. 6A do not have to approximate the characteristic impedance of theintra-hybrid connection lines in order to appear transparent. This maymake it easier to implement the tuning circuit 24B. Also, since thefirst RF filter 106 and the second RF filter 108 are coupled in shunt,the first RF filter 106 and the second RF filter 108 can provide pathsfor neutralizing parasitic loading effects within the RF receive bandusing reactive compensation.

By placing the passband in the RF receive band, the RF filter circuit22B passes the first RF QHRS 56 and the second RF QHRS 58 to the secondhybrid coupler 20′. In this particular embodiment, the first RF filter106 passes the first RF QHRS 56 to the second hybrid coupler 20′, whilethe second RF filter 108 passes the second RF QHRS 58 to the secondhybrid coupler 20′. More specifically, the first RF QHRS 56 is receivedat the fifth port 84 of the second hybrid coupler 20′, and the second RFQHRS 58 is received at the sixth port 86 of the second hybrid coupler20′.

As described above with respect to FIG. 3A, the second hybrid coupler20′ shown in FIG. 3A is configured to provide a quadrature phase shiftfrom the sixth port 86 to the seventh port 88. In this example, thephase shift is 90 degrees (or π/2 radians), and thus the second RF QHRS58 has a phase, as seen from the seventh port 88, of 180 degrees (notethat the second RF QHRS 58 was received with a phase of 90 degrees inthis example, and thus is seen with a phase of 180 degrees with anotherphase shift of 90 degrees). No phase shift is provided from the fifthport 84 to the eighth port 90. Thus, the phase difference between thefirst RF QHRS 56 and the second RF QHRS 58 as seen from the seventh port88 is about 180 degrees (note that the first RF QHRS 56 was receivedwith a phase of zero (0) degrees and thus is seen with a phase of zero(0) degrees at the seventh port 88). Accordingly, the quadrature phaseshift at the seventh port 88 from the sixth port 86 results indestructive interference between the first RF QHRS 56 and the second RFQHRS 58 at the seventh port 88. As a result, the first RF QHRS 56 andthe second RF QHRS 58 are substantially cancelled at the seventh port88. In this manner, the seventh port 88 is substantially isolated fromreceive signal flow.

The second hybrid coupler 20′ is configured to output the RF receiveoutput signal 30 from the eighth port 90 in response to the first RFQHRS 56 being received from the RF filter circuit 22B at the fifth port84 and the second RF QHRS 58 being received from the RF filter circuit22B at the sixth port 86. In this particular embodiment, the secondhybrid coupler 20′ is configured to pass the second RF QHRS 58 from thesixth port 86 to the eighth port 90. The second hybrid coupler 20′provides no phase shift to the second RF QHRS 58 from the sixth port 86to the eighth port 90. Since no phase shift is provided to the second RFQHRS 58 from the sixth port 86 to the eighth port 90, the second RF QHRS58 is thus passed with a phase of 90 degrees (or π/2 radians) to theeighth port 90. The second hybrid coupler 20′ is configured to provide aquadrature phase shift to the first RF QHRS 56 at the eighth port 90.Thus, since the quadrature phase shift is provided to the first RF QHRS56 from the fifth port 84 to the eighth port 90, the second RF QHRS 58is passed with a phase of 90 degrees (or π/2 radians) to the eighth port90.

Accordingly, the first RF QHRS 56 is provided substantially as aduplicate of the second RF QHRS 58 at the eighth port 90. As a result,the first RF QHRS 56 and the second RF QHRS 58 constructively interfereat the eighth port 90 to output the RF receive output signal 30 from theeighth port 90. Note that since the first RF QHRS 56 and the second RFQHRS 58 substantially cancel at the seventh port 88 due to destructiveinterference, very little or no power is transferred from the first RFQHRS 56 and the second RF QHRS 58 to the seventh port 88. Instead most,if not all, of the power in the first RF QHRS 56 and the second RF QHRS58 is transferred to the eighth port 90 and provided in the RF receiveoutput signal 30.

The impedance load 100 is coupled to the seventh port 88 of the secondhybrid coupler 20′. The impedance load 100 may be a 50 Ohm load. Due tothe phase-shifting provided by the first hybrid coupler 18′ and thesecond hybrid coupler 20′, spurious emissions from the second port 78 ofthe first hybrid coupler 18 would see a very high (theoreticallyinfinite) impedance level at the eighth port 90 of the second hybridcoupler, but only the impedance load 100 at the seventh port 88. Thus,the spurious emissions are aggregated to be the aggregated noise signal102 at the seventh port 88. The aggregated noise signal 102 isdissipated by the impedance load 100. Additionally, the eighth port 90is isolated from the seventh port 88. As such, the seventh port 88 issubstantially unresponsive to signals incident at the eighth port 90 andthe eighth port 90 is substantially unresponsive to signals incident atthe seventh port 88.

FIG. 6B illustrates the tunable RF duplexer 10B along with thetransmission signal flow. The first hybrid coupler 18′ is operable toreceive the RF transmission input signal 26 at the second port 78. TheRF transmission input signal 26 operates in the RF transmission band,which is within the same RF communication band as the RF receive band ofthe RF receive input signal 14 (see FIG. 2A). The first hybrid coupler18′ is configured to split the RF transmission input signal 26 into thefirst RF QHTS 60 and the second RF QHTS 62. Since the first RF QHTS 60and the second RF QHTS 62 are quadrature hybrids, the first RF QHTS 60and the second RF QHTS 62 are approximately equal in power, but have aquadrature phase difference of 90 degrees or π/2 radians. The firsthybrid coupler 18′ is operable to output the first RF QHTS 60 from thefourth port 82 and to output the second RF QHTS 62 from the third port80 in response to receiving the RF transmission input signal 26 at thesecond port 78.

The RF filter circuit 22B is tunable so as to reflect the first RF QHTS60 and the second RF QHTS 62 back to the first hybrid coupler 18′. Thefrequency response of the RF filter circuit 22B defines a stopband andthe RF filter circuit 22B is tunable so as to shift the stopband. Forexample, the stopband may be a notch that is shiftable. The tuningcircuit 24B is configured to tune the frequency response of the RFfilter circuit 22B so that the signal bandwidth of the first RF QHTS 60and the signal bandwidth of the second RF QHTS 62 are each within thestopband of the RF filter circuit 22B. In this embodiment, the tuningcircuit 24B is configured to tune a stopband of the second RF filter 108within the signal bandwidth of the first RF QHTS 60 and to tune astopband of the first RF filter 106 within the signal bandwidth of thesecond RF QHTS 62. To provide each of the stopbands, the first RF filter106 and the second RF filter 108 each appear approximately as shortcircuits within the RF transmission band.

In this embodiment, the frequency response of the RF filter circuit 22Bdefines the stopband of the RF filter circuit 22B as a notch. The tuningcircuit 24B may be configured to place the notch within the RFtransmission band so that the notch is centered at the RF transmissionsignal frequency. More specifically, the frequency response of the firstRF filter 106 defines its stopband as a notch. Similarly, the frequencyresponse of the second RF filter 108 is a notch. The tuning circuit 24Bis configured to generate the transmission tuning control output 112such that variable reactive impedance components in both the first RFfilter 106 and the second RF filter 108 are responsive to one or moresignal levels of the transmission tuning control output 112 and adjustthe variable impedance levels based on the one or more signal levels ofthe transmission tuning control output 112. As a result, the notchdefined by the individual frequency response of the first RF filter 106is shifted to include the signal bandwidth of the second RF QHTS 62.Also, the notch defined by the individual frequency response of thesecond RF filter 108 is shifted to include the signal bandwidth of thefirst RF QHTS 60. In other words, the notches defined by the individualfrequency responses of the first RF filter 106 and the second RF filter108 are placed within the RF transmission band.

Since the first RF filter 106 is coupled in shunt and operatesapproximately as a short in the RF transmission band, the first RFfilter 106 reflects the second RF QHTS 62 back to the third port 80 ofthe first hybrid coupler 18′. Also, since the second RF filter 108 iscoupled in shunt and operates approximately as a short in the RFtransmission band, the second RF filter 108 reflects the first RF QHTS60 back to the fourth port 82 of the first hybrid coupler 18′.Accordingly, the second hybrid coupler 20′ is substantially isolatedfrom the transmission signal flow. The first hybrid coupler 18′ isconfigured to combine the first RF QHTS 60 and the second RF QHTS 62into the RF transmission output signal 16. To combine the first RF QHTS60 and the second RF QHTS 62 into the RF transmission output signal 16,the first hybrid coupler 18′ is configured to pass the second RF QHTS 62from the third port 80 to the first port 76. Additionally, the firsthybrid coupler 18′ is configured to pass the first RF QHTS 60 from thefourth port 82 to the first port 76. However, the first hybrid coupler18′ provides a quadrature phase shift to the first RF QHTS 60 from thefourth port 82 to the first port 76. Thus, the first RF QHTS 60 isprovided substantially as a duplicate of the second RF QHTS 62 at thefirst port 76. For example, if the phase of the second RF QHTS 62 is 90degrees at the third port 80, the second RF QHTS 62 has a phase of 90degrees at the first port 76. Additionally, the phase of the first RFQHTS 60 at the fourth port 82 is approximately zero (0) degrees.However, due to the quadrature phase shift between the fourth port 82and the first port 76, the first RF QHTS 60 has a phase of about 90degrees at the first port 76. Accordingly, the first RF QHTS 60 and thesecond RF QHTS 62 constructively interfere at the first port 76 tooutput the RF transmission output signal 16 from the first port 76.

Also, note that the first hybrid coupler 18′ is configured such that thequadrature phase shift at the second port 78 results in destructiveinterference between the first RF QHTS 60 and the second RF QHTS 62.Referring again to the previous example provided, at the second port 78,the first RF QHTS 60 appears to have a phase of zero degrees, but thesecond RF QHTS 62 appears to have a phase of 180 degrees. As a result,the first RF QHTS 60 and the second RF QHTS 62 are substantiallycancelled at the second port 78. Consequently, most, if not all, of thepower of the first RF QHTS 60 and the second RF QHTS 62 is transferredto the first port 76 and provided in the RF transmission output signal16. The first hybrid coupler 18′ is thus configured to output the RFtransmission output signal 16 from the first port 76 in response to thefirst RF QHTS 60 being reflected back by the RF filter circuit 22B tothe fourth port 82 and the second RF QHTS 62 being reflected back by theRF filter circuit 22B to the third port 80.

Note that spurious emissions from the first RF QHTS 60 and the second RFQHTS 62 may not be reflected by the RF filter circuit 22B. Instead,spurious noise emissions outside the stopband, may be passed toward thesecond hybrid coupler 20′. In this example, the spurious emissions woulddestructively interfere at the eighth port 90 and constructivelyinterfere at the seventh port 88. Accordingly, these spurious emissionsbecome part of the aggregated noise signal 102 and are dissipated by theimpedance load 100. Therefore, the tunable RF duplexer 10B significantlyreduces noise interference from the second port 78 of the first hybridcoupler 18′ (see FIG. 3A) at the eighth port 90.

As shown in FIGS. 6A and 6B, the tunable RF duplexer 10B allows thereceive signal flow and the transmission signal flow to be simultaneous.More specifically, the first port 76 of the first hybrid coupler 18′ isconfigured to simultaneously receive the RF receive input signal 14 andoutput the RF transmission output signal 16 to and from the antenna 12.As a result, the RF transmission output signal 16 is output from thefirst port 76 while the RF receive input signal 14 is beingsimultaneously received at the first port 76. The tunable RF duplexer10B can provide this functionality because the second port 78 issubstantially isolated from the eighth port 90.

Referring now to FIG. 7, FIG. 7 illustrates one embodiment of afrequency response 48B provided by the RF filter circuit 22B shown inFIGS. 6A and 6B. The frequency response 48B defines a passband 50B and astopband 52B. The RF filter circuit 22B described in FIGS. 6A and 6B istunable so that the RF communication band (referred to generically aselement 34, and specifically as elements 34A-34E) can be switched to beany one of the plurality of different RF communication bands 34A-34E.The different RF communication bands 34A-34E may each define differentRF receive band(s) and different RF transmission band(s) within each ofthe RF communication bands 34A-34E. Accordingly, the RF transmissioninput signal 26 (shown in FIGS. 6A-6B) and the RF receive input signal14 (shown in FIGS. 6A-6B) may be in the respective RF receive band andRF transmission band of any of the different RF communication bands34A-34E. Accordingly, the first RF QHRS 56, the second RF QHRS 58, andthe RF receive output signal 30 would also operate within the RF receiveband of the RF receive input signal 14. Similarly, the first RF QHTS 60,the second RF QHTS 62, and the RF transmission output signal 16 wouldoperate in the respective RF transmission band of the RF transmissioninput signal 26.

When the RF communication band 34 is switched to be a different one ofthe plurality of RF communication bands 34A-34E, the tuning circuit 24Bis operable to tune the frequency response 48B to the particular RFcommunication band 34A-34E. Since the RF communication band 34 has beenswitched, the first RF QHRS 56 and the second RF QHRS 58 are provided inthe new RF communication band 34A-34E. In particular, the first RF QHRS56 and the second RF QHRS 58 operate within the RF receive band of thenew RF communication band 34A-34E. Also, since the RF communication band34 has been switched to the new RF communication band 34A-34E, the firstRF QHTS 60 and the second RF QHTS 62 are provided within the new RFcommunication band 34A-34E. In particular, the first RF QHTS 60 and thesecond RF QHTS 62 operate within the RF transmission band of the new RFcommunication band 34A-34E. The tuning circuit 24B tunes the passband50B so as to pass the first RF QHRS 56 and the second RF QHRS 58 inresponse to the RF communication band 34 being switched to the new RFcommunication band 34A-34E. More particularly, the passband 50B isprovided so as to include the RF receive band of the selected RFcommunication band 34A-34E. Similarly, the tuning circuit 24B shifts thestopband 52B to reflect the first RF QHTS 60 and the second RF QHTS 62in response to the RF communication band 34 being switched to the new RFcommunication band 34A-34E. As such, the stopband 52B is shifted toinclude the RF transmission band of the selected RF communication band34A-34E.

Referring to FIGS. 6A, 6B, and 8A, FIG. 8A illustrates one embodiment ofa tunable hybrid coupler 114. In practice, the tunable RF duplexer 10Bshown in FIGS. 6A and 6B may not be able to provide infinitetransmission/receive isolation. For example, impedance mismatchesbetween the first hybrid coupler 18′ and the second hybrid coupler 20′may result in small reflections and leaks that degradetransmission/receive isolation. To help reduce and/or eliminate thesereflections and leaks, the first hybrid coupler 18′ and the secondhybrid coupler 20′ are tunable. As mentioned above, the first hybridcoupler 18′ and the second hybrid coupler 20′ in FIGS. 6A and 6B mayeach be tunable. Accordingly, in one exemplary embodiment of the tunableRF duplexer 10B, the first hybrid coupler 18′ and the second hybridcoupler 20′ are each provided as the tunable hybrid coupler 114 of FIG.8A.

The tunable hybrid coupler 114 shown in FIG. 8A is a lumped-elementhybrid coupler. However, the tunable hybrid coupler 114 may beimplemented using any suitable hybrid coupler topology. In alternativeembodiments, the tunable hybrid coupler 114 may be a rat-race hybridcoupler, transmission line coupler, Wilkinson hybrid coupler, striplinehybrid coupler, microstrip hybrid coupler, ferrite core hybrid coupler,amalgamations of different types of hybrid couplers, and/or the like.The tunable hybrid coupler 114 has a first port 116, a second port 118,a third port 120, and a fourth port 122. With regard to the first hybridcoupler 18′ shown in FIGS. 6A-6B, the first port 116 of the tunablehybrid coupler 114 corresponds to the first port 76 of the first hybridcoupler 18′, the second port 118 of the tunable hybrid coupler 114corresponds to the second port 78 of the first hybrid coupler 18′, thethird port 120 of the tunable hybrid coupler 114 corresponds to thethird port 80 of the first hybrid coupler 18′, and the fourth port 122of the tunable hybrid coupler 114 corresponds to the fourth port 82 ofthe first hybrid coupler 18′. With regard to the second hybrid coupler20′ shown in FIGS. 6A-6B, the first port 116 of the tunable hybridcoupler 114 corresponds to the fifth port 84 of the second hybridcoupler 20′, the second port 118 of the tunable hybrid coupler 114corresponds to the sixth port 86 of the second hybrid coupler 20′, thethird port 120 of the tunable hybrid coupler 114 corresponds to theseventh port 88 of the second hybrid coupler 20′, and the fourth port122 of the tunable hybrid coupler 114 corresponds to the eighth port 90of the second hybrid coupler 20′.

The tunable hybrid coupler 114 shown in FIG. 8A includes a firstinductive element 124 and a second inductive element 126. The firstinductive element 124 is connected in series between the fourth port 122and the second port 118. The second inductive element is connected inseries between the third port 120 and the first port 116. In thisembodiment, the first inductive element 124 and the second inductiveelement 126 are inductively coupled and have a coupling coefficient kthat is equal to approximately one (1). As such, the first inductiveelement 124 is operable to generate a magnetic flux in response to acurrent propagating on the first inductive element 124. The magneticflux generated by the first inductive element 124 also flows within thesecond inductive element 126. The magnetic flux thus induces anelectromotive force on the second inductive element 126 so as togenerate a current that propagates on the second inductive element 126.However, a quadrature phase shift of 90 degrees or π/2 radians isprovided from the first inductive element 124 to the second inductiveelement 126. The current that propagates on the second inductive element126 has a phase of 90 degrees or π/2 radians with respect to the currentthat propagates through the first inductive element 124. Since thecoupling coefficient k between the first inductive element 124 and thesecond inductive element 126 is one (1), the currents propagating on thefirst inductive element 124 and the second inductive element 126 haveapproximately the same magnitude. In other words, half of the power(ignoring parasitic and non-ideal circuit behavior) is transferredthrough the magnetic flux from the first inductive element 124 and thesecond inductive element 126.

Similarly, the second inductive element 126 is operable to generate amagnetic flux in response to a current propagating on the secondinductive element 126. The magnetic flux generated by the secondinductive element 126 also flows within the first inductive element 124.The magnetic flux thus induces an electromotive force on the firstinductive element 124 so as to generate a current that propagates on thefirst inductive element 124. However, a quadrature phase shift of 90degrees or π/2 radians is provided from the second inductive element 126to the first inductive element 124. The current that propagates on thefirst inductive element 124 has a phase of 90 degrees or π/2 radianswith respect to the current that propagates through the second inductiveelement 126. Since the coupling coefficient k between the secondinductive element 126 and the first inductive element 124 is one (1),the currents propagating on the second inductive element 126 and thefirst inductive element 124 have approximately the same magnitude. Inother words, half of the power (ignoring parasitic and non-ideal circuitbehavior) is transferred through the magnetic flux from the secondinductive element 126 to the first inductive element 124. In thisembodiment, the first inductive element 124 has a first inductance andthe second inductive element 126 has a second inductance, wherein thefirst inductance and the second inductance are the same.

Accordingly, the tunable hybrid coupler 114 is configured to split an RFsignal received at one of the ports 116, 118, 120, 122 into a pair of RFquadrature hybrid signals (QHS). One of the RF QHS generated from thesplitting of the RF signal propagates through the inductive element124,126 in series with the port 116, 118, 120, 122 that received the RFsignal. The tunable hybrid coupler 114 is configured to output this RFQHS from the other port 120, 122, 116, 118 in series with the inductiveelement 124, 126. In this embodiment, the RF QHS output from the otherport 120, 122, 116, 118 in series with the inductive element 124, 126 isphase-aligned with the RF signal. However, in alternative embodiments, aphase shift of Δ (i.e., such as +45 degrees or +π/4 radians) may beprovided from the port 116, 118, 120, 122 that receives the RF signaland the other port 120, 122, 116, 118 in series with the inductiveelement 124, 126.

The other RF QHS generated from the splitting of the RF signalpropagates through the other inductive element 126,124 that isinductively coupled with the inductive element 124,126. Each RF QHS hashalf the spectral power density of the RF signal received at the port116, 118, 120, 122. The other RF QHS propagates in the same direction asthe RF signal. Thus, the tunable hybrid coupler 114 is configured tooutput the other RF QHS propagating on the other inductive element126,124 at the port 122, 120, 118, 116. In this embodiment, the other RFQHS output from the port 122, 120, 118, 116 has a phase of +90 degreesor +π/2 radians with respect to the RF signal. However, in alternativeembodiments, a phase shift of Δ±90 degrees (such as +135 degrees or+3π/4 radians, −45 degrees or −π/4 radians) may be provided from theport 116, 118, 120, 122 that receives the RF signal and the port 122,120, 118, 116 that outputs the other RF QHS. Furthermore, note that thatthe port 118, 116, 122, 120 is isolated from the port 116, 118, 120, 122that received the RF signal. As mentioned above, the first hybridcoupler 18′ and the second hybrid coupler 20′ of FIGS. 6A and 6B mayeach be implemented as the tunable hybrid coupler 114 shown in FIG. 8A.In this manner, the tunable RF duplexer 10B can simultaneously receivethe RF receive input signal 14 and the RF transmission input signal 26,generate the first RF QHRS 56, the second RF QHRS 58, the first RF QHTS60, and the second RF QHTS 62, and output the RF receive output signal30 and the RF transmission output signal 16, as described above withrespect to FIGS. 6A and 6B.

Referring to FIG. 8A, the tunable hybrid coupler 114 includes a variablecapacitive element 128 and a variable capacitive element 130. Thevariable capacitive element 128 is connected between the first port 116and the second port 118. The variable capacitive element 128 is operableto provide a first variable capacitance. As a result, a first variablecapacitive impedance is presented by the variable capacitive element 128between the first port 116 and the second port 118. In addition, thevariable capacitive element 130 is coupled between the third port 120and the fourth port 122. The variable capacitive element 130 is operableto provide a second variable capacitance. As a result, a second variablecapacitive impedance is presented by the variable capacitive element 130between the third port 120 and the fourth port 122.

Accordingly, the first inductive element 124, the second inductiveelement 126, the variable capacitive element 128, and the variablecapacitive element 130 form an impedance matching network for a 4-portsystem. The impedance matching network provides impedancetransformations so as to increase impedance matching between input andoutput impedances at the first port 116, the second port 118, the thirdport 120, and the fourth port 122. The term “input impedance” refers toan internal impedance at the respective port 116, 118, 120, 122 towardthe tunable hybrid coupler 114. “Output impedance” refers to an externalimpedance at the respective port 116, 118, 120, 122 out of the tunablehybrid coupler 114. The impedance matching network also provides animpedance transformation so as to increase impedance matching between animpedance from the first port 116 to the second port 118 and animpedance from the third port 120 to the fourth port 122.

In this embodiment, the impedance matching network provides an impedancetransformation that increases impedance matching between an inputimpedance and an output impedance at the first port 116. Additionally,the impedance matching network provides an impedance transformation thatincreases impedance matching between an input impedance and an outputimpedance at the second port 118. The impedance matching network alsoprovides an impedance transformation that increases impedance matchingbetween an input impedance and an output impedance at the third port120. Finally, the impedance matching network provides an impedancetransformation that increases impedance matching between an inputimpedance and an output impedance at the fourth port 122.

As shown in FIG. 8A, the tunable hybrid coupler 114 also includes avariable capacitive element 132 and a variable capacitive element 134.The variable capacitive element 132 is connected between the first port116 and the third port 120 in parallel with the first inductive element124. The variable capacitive element 132 is operable to provide a thirdvariable capacitance. As a result, a third variable capacitive impedanceis presented by the variable capacitive element 132 between the firstport 116 and the third port 120 in parallel with the first inductiveelement 124. In addition, the variable capacitive element 134 is coupledbetween the second port 118 and the fourth port 122 in parallel with theinductive element 126. The variable capacitive element 134 is operableto provide a fourth variable capacitance. As a result, a fourth variablecapacitive impedance is presented by the variable capacitive element 134between the second port 118 and the fourth port 122 in parallel with theinductive element 126. The variable capacitive element 132 and thevariable capacitive element 134 allow for more accurate tuning of theimpedance transformations. Additionally, the variable capacitive element132 in parallel with the first inductive element 124 can be used todefine a notch. Similarly, the variable capacitive element 134 inparallel with the second inductive element 136 can also be used todefine a notch.

Accordingly, the variable capacitive element 128, the variablecapacitive element 130, the variable capacitive element 132, and thevariable capacitive element 134 allow the tunable hybrid coupler 114 tobe tuned. In this manner, the tunable hybrid coupler 114 is operable toprovide impedance transformations to increase impedance matching of theinput and output impedances at the first port 116, the second port 118,the third port 120, and the fourth port 122 within different RFcommunication bands. Ideally, the tunable hybrid coupler 114 perfectlymatches the input impedances and the output impedances at each of theports 116, 118,120, 122. However, in practice, there may be somemismatches between input and output impedances despite the impedancetransformations. An acceptable amount of mismatch may depend on theparticular application for the tunable hybrid coupler 114 along with RFcommunication band specifications. For example, the tunable hybridcoupler 114 shown in FIG. 8A is sufficiently broadband such that theimpedance transformations increase matching of the input and outputimpedances both within the transmission frequency band and the receivefrequency band of a selected RF communication band. However, for anyselected capacitance value of the variable capacitances, the tunablehybrid coupler 114 may not be capable of providing perfect matchingwithin both the transmission frequency band and the receive frequencyband and/or the transmission frequency and the receive frequencysimultaneously. Therefore, in some circumstances, the variablecapacitances may be selected at an interpolated point that does notprovide perfect matching within or at either the transmission frequencyband/transmission frequency or the receive frequency band/receivefrequency.

The tunable hybrid coupler 114 is configured to received the hybridcontrol output 111 (in FIGS. 6A-6B, the first hybrid coupler 18′ isoperable to receive the first hybrid control output 111A and the secondhybrid coupler 20′ is operable to receive the second hybrid controloutput 111B). The hybrid control output 111 includes one or more hybridcontrol signals 111-1, 111-2, 111-3, and 111-4. Each of the variablecapacitive elements 128, 130, 132, 134 is configured to set itsrespective variable capacitance in accordance with a signal level of acorresponding one of the hybrid control signals 111-1, 111-2, 111-3, and111-4. For example, the variable capacitive element 128 is configured toset the first variable capacitance in accordance with a signal level ofthe hybrid control signal 111-1. Additionally, the variable capacitiveelement 130 is configured to set the second variable capacitance inaccordance with a signal level of the hybrid control signal 111-2.Furthermore, the variable capacitive element 132 is configured to setthe third variable capacitance in accordance with a signal level of thehybrid control signal 111-3. Finally, the variable capacitive element134 is configured to set the third variable capacitance in accordancewith a signal level of the hybrid control signal 111-4. Thus, the firstvariable capacitance, the second variable capacitance, the thirdvariable capacitance, and the fourth variable capacitance are eachconfigured to be adjusted in accordance with the respective signal levelof the corresponding one of the respective hybrid control signals 111-1,111-2, 111-3, and 111-4. In this manner, the tunable hybrid coupler 114can be tuned to increase impedance matching between input and outputimpedances at the ports 116, 118,120, 122 within the different RFcommunication bands.

FIG. 8B illustrates one embodiment of the tunable hybrid coupler 114formed on a semiconductor substrate 136. The semiconductor substrate 136has a substrate body 138 formed from a wafer and/or doped layers of asuitable semiconductor material. For example, the semiconductor materialmay be Silicon (Si), Silicon Germanium (SiGe), Gallium Arsenide (GaAs),Indium Phosphorus (InP), and/or the like. Typical dopants that may beutilized to dope the semiconductor layers are Gallium (Ga), Arsenic(As), Silicon (Si), Tellurium (Te), Zinc (Zn), Sulfur (S), Boron (B),Phosphorus (P), Aluminum Gallium Arsenide (AlGaAs), Indium GalliumArsenide (InGaAs), and/or the like. Furthermore, metallic layers,insulating layers, and the like may be formed on one or more surfaces orwithin the substrate body 138 to provide terminals, traces, coils,connections, passive impedance elements, active semiconductorcomponents, and/or the like. Also, any type of suitable semiconductortechnology may be used to provide the topology of the semiconductorsubstrate 136. For example, the semiconductor technology may beComplementary Metal-On-Oxide Semiconductor technology (CMOS),BiComplementary Metal-On-Oxide Semiconductor technology (BiCMOS),Silicon-On-Insulator technology (SOI), and/or the like. In thisembodiment, the semiconductor technology of the semiconductor substrate136 is SOI, and thus the semiconductor material of the substrate body138 is Si. Additionally, the substrate body 138 has a surface 140. Thesurface 140 thus defines a direction 142 normal to the surface 140.Terms in this disclosure referring to directional words such as “over”or “top” and “under” or “beneath” are made with respect to the direction142 defined by the surface 140 of the substrate body 138. It should benoted that the first hybrid coupler 18′, the second hybrid coupler 20′,the first RF filter 106, and the second RF filter 108 shown in FIGS. 6Aand 6B may be formed with the same semiconductor substrate 136, or oneor more of these components may be formed on one or more separatesemiconductor substrates.

As shown in FIG. 8B, the first port 116, the second port 118, the thirdport 120, and the fourth port 122 are each formed as a terminal on thesurface 140 of the substrate body 138. In this manner, RF signals may beinput and/or output from the tunable hybrid coupler 114. The firstinductive element 124 is formed as a spiral coil on the surface 140. Thefirst inductive element 124 and the first port 116, the second port 118,the third port 120, and the fourth port 122 may be formed from ametallic layer on the surface 140 that has been etched to provide thefirst inductive element 124 and the first port 116, the second port 118,the third port 120, and the fourth port 122. At a center of the spiralcoil, the first inductive element 124 shown in FIG. 8B defines a centralaxis 144 approximately parallel to the direction 142 and thus normal tothe surface 140. The first inductive element 124 is wound about thecentral axis 144 to form turns that have increasing circumferences themore outwardly the turns are from the central axis 144. Additionally,the second inductive element 126 is formed as a spiral coil within thesubstrate body 138. The second inductive element 126 may be formed froma metallic layer within the substrate body 138. The metallic layer maybe formed on a layer beneath the surface 140 and etched to provide thesecond inductive element 126. At a center of the spiral coil, the secondinductive element 126 shown in FIG. 8B defines a central axis 146approximately parallel to the direction 142 and thus normal to thesurface 140. The second inductive element 126 is wound about the centralaxis 146 to form turns that have increasing circumferences the moreoutwardly the turns are from the central axis 146. The first inductiveelement 124 and the second inductive element 126 are inductively coupledso that the magnetic flux of each one of the inductive elements 124, 126induces a current in the other one of the inductive elements 126, 124.

FIG. 8B also illustrates the variable capacitive elements 128, 130, 132,and 134, which are formed as variable metal-insulator-metal capacitorsin the semiconductor substrate 136. However, with the arrangement shownin FIG. 8B, the first inductive element 124 and the second inductiveelement 126 are configured so that there is a parasitic capacitancebetween the first inductive element 124 and the second inductive element126. In this embodiment, the parasitic capacitance between the firstinductive element 124 and the second inductive element 126 issignificantly greater than a highest capacitive value of the variablecapacitances provided by the variable capacitive elements 128, 130, 132,and 134. As such, the largest and greatest capacitive impedance is fromthis parasitic capacitance between the inductive elements 126, 124.Also, the parasitic capacitance sets a minimum capacitance between thefirst inductive element 124 and the second inductive element 126. Whilethe variable capacitance of the variable capacitive element 128 can beused to adjust a capacitive impedance as seen between the first port 116and the second port 118 and the variable capacitance of the variablecapacitive element 130 can be used to adjust a capacitive impedance asseen between the third port 120 and the fourth port 122, the primarycapacitive impedance between the first port 116 and the second port 118and between the third port 120 and the fourth port 122 is from theparasitic capacitance between the first inductive element 124 and thesecond inductive element 126.

As such, the parasitic capacitance between the first inductive element124 and the second inductive element 126 sets a minimum capacitance ofthe tunable hybrid coupler 114. Note that in this embodiment, there is adisplacement along the surface 140 from the central axis 144 of thefirst inductive element 124 to the central axis 146 of the secondinductive element 126. Since the first inductive element 124 and thesecond inductive element 126 are approximately parallel, thisdisplacement sets the parasitic capacitance and the minimum capacitanceof the tunable hybrid coupler 114. Accordingly, when manufacturing thetunable hybrid coupler 114, this displacement may be selected to set theparasitic capacitance in accordance with a frequency range of thedifferent RF communication bands for a particular design application.During operation, adjustments can be made to the variable capacitancesof the variable capacitive elements 128, 130, 132, and 134 so as to tunethe tunable hybrid coupler 114 to a selected one of the different RFcommunication bands.

FIG. 9A illustrates another embodiment of a tunable hybrid coupler 114′.Like the tunable hybrid coupler 114 shown in FIG. 8A, the tunable hybridcoupler 114′ also has the first port 116, the second port 118, the thirdport 120, and the fourth port 122. Similarly, with regard to the firsthybrid coupler 18′ shown in FIGS. 6A-6B, the first port 116 of thetunable hybrid coupler 114′ corresponds to the first port 76 of thefirst hybrid coupler 18′, the second port 118 of the tunable hybridcoupler 114′ corresponds to the second port 78 of the first hybridcoupler 18′, the third port 120 of the tunable hybrid coupler 114′corresponds to the third port 80 of the first hybrid coupler 18′, andthe fourth port 122 of the tunable hybrid coupler 114′ corresponds tothe fourth port 82 of the first hybrid coupler 18′. With regard to thesecond hybrid coupler 20′ shown in FIGS. 6A-6B, the first port 116 ofthe tunable hybrid coupler 114′ corresponds to the fifth port 84 of thesecond hybrid coupler 20′, the second port 118 of the tunable hybridcoupler 114′ corresponds to the sixth port 86 of the second hybridcoupler 20′, the third port 120 of the tunable hybrid coupler 114′corresponds to the seventh port 88 of the second hybrid coupler 20′, andthe fourth port 122 of the tunable hybrid coupler 114′ corresponds tothe eighth port 90 of the second hybrid coupler 20′.

Also like the tunable hybrid coupler 114 shown in FIG. 8A, the tunablehybrid coupler 114′ includes the first inductive element 124 connectedin series between the fourth port 122 and the second port 118 andincludes the second inductive element 126 connected in series betweenthe third port 120 and the first port 116. The first inductive element124 and the second inductive element 126 are inductively coupled andhave the coupling coefficient k that is equal to approximately one (1).

Referring to FIG. 9A, the tunable hybrid coupler 114′ includes avariable capacitive element 128′, a variable capacitive element 130′, avariable capacitive element 132′, a variable capacitive element 134′, avariable capacitive element 148, and a variable capacitive element 150.The variable capacitive element 128′ is connected between the first port116 and the second port 118 so as to provide a first variablecapacitance between the first port 116 and the second port 118.

As with the tunable hybrid coupler 114 illustrated in FIG. 8A, the firstinductive element 124, the second inductive element 126, the variablecapacitive element 128′, and the variable capacitive element 130′ shownin FIG. 9A form an impedance matching network for a 4-port system. Theimpedance matching network provides impedance transformations so as toincrease impedance matching between input and output impedances at thefirst port 116, the second port 118, the third port 120, and the fourthport 122. Accordingly, the impedance matching network provides animpedance transformation that increases impedance matching between aninput impedance and an output impedance at the first port 116.Additionally, the impedance matching network provides an impedancetransformation that increases impedance matching between an inputimpedance and an output impedance at the second port 118. The impedancematching network also provides an impedance transformation thatincreases impedance matching between an input impedance and an outputimpedance at the third port 120. Finally, the impedance matching networkprovides an impedance transformation that increases impedance matchingbetween an input impedance and an output impedance at the fourth port122.

As shown in FIG. 9A, the tunable hybrid coupler 114′ also includes thevariable capacitive element 132′ and the variable capacitive element134′. The variable capacitive element 132′ is connected between thefirst port 116 and the third port 120 in parallel with the firstinductive element 124. The variable capacitive element 132′ is operableto provide a third variable capacitance. As a result, a third variablecapacitive impedance is presented by the variable capacitive element132′ between the first port 116 and the third port 120 in parallel withthe first inductive element 124. In addition, the variable capacitiveelement 134′ is coupled between the second port 118 and the fourth port122 in parallel with the second inductive element 126. The variablecapacitive element 134′ is operable to provide a fourth variablecapacitance. As a result, a fourth variable capacitive impedance ispresented by the variable capacitive element 134′ between the secondport 118 and the fourth port 122 in parallel with the second inductiveelement 126. The variable capacitive element 132′ and the variablecapacitive element 134′ allow for more accurate tuning of the impedancetransformations. Additionally, the variable capacitive element 132′ inparallel with the first inductive element 124 can be used to define anotch. Similarly, the variable capacitive element 134′ in parallel withthe second inductive element 126 can be also used to define a notch.

However, in order to have strong inductive coupling between the firstinductive element 124 and the second inductive element 126, the firstinductive element 124 and the second inductive element 126 may haverelatively large inductances for any given minimum capacitance. Thelarge inductances of the first inductive element 124 and the secondinductive element 126 could restrict a frequency range for tuning thenotches. In order to expand the frequency range for the notches, thetunable hybrid coupler 114′ also includes the variable capacitiveelement 148 and the variable capacitive element 150. The variablecapacitive element 148 is connected in parallel with a portion 152 ofthe first inductive element 124. More specifically, the variablecapacitive element 148 of FIG. 9A is connected from the first port 116to an intermediary node 154 of the first inductive element 124. Thevariable capacitive element 148 is operable to provide a fifth variablecapacitance. As such, unlike the variable capacitive element 132′, thevariable capacitive element 148 does not have to resonate with theentire first inductive element 124 to create a notch. Rather, thevariable capacitive element 148 is configured to create the notch byresonating with only the portion 152 of the first inductive element 124.Since the portion 152 of the first inductive element 124 has a smallerinductance than the entire first inductive element 124, the notchcreated by the portion 152 of the first inductive element 124 and thevariable capacitive element 148 can be tuned to a greater frequencyrange than the frequency range of the notch created by the variablecapacitive element 132′ and the entire first inductive element 124.

Similarly, the variable capacitive element 150 is connected in parallelwith a portion 156 of the second inductive element 126. Morespecifically, the variable capacitive element 150 of FIG. 9A isconnected from the second port 118 to an intermediary node 158 of thesecond inductive element 126. The variable capacitive element 150 isoperable to provide a sixth variable capacitance. As such, unlike thevariable capacitive element 134′, the variable capacitive element 150does not have to resonate with the entire second inductive element 126to create a notch. Rather, the variable capacitive element 150 isconfigured to create the notch by resonating with only the portion 156of the second inductive element 126. Since the portion 156 of the secondinductive element 126 has a smaller inductance than the entire secondinductive element 126, the notch created by the portion 156 of thesecond inductive element 126 and the variable capacitive element 150 canbe tuned to a greater frequency range than the frequency range of thenotch created by the variable capacitive element 134′ and the entiresecond inductive element 126.

The tunable hybrid coupler 114′ is configured to receive the hybridcontrol output 111 (in FIGS. 6A-6B, the first hybrid coupler 18′ isoperable to receive the first hybrid control output 111A and the secondhybrid coupler 20′ is operable to receive the second hybrid controloutput 111B). However, in FIG. 9A, each of the variable capacitiveelements 128′, 130′, 132′, 134′, 148, and 150 is a programmablecapacitor array (PCA). As such, each one of the variable capacitiveelements 128′, 130′, 132′, 134′, 148, and 150 includes a group ofselectable capacitive components. These selectable capacitive componentsin each of the variable capacitive elements 128′, 130′, 132′, 134′, 148,and 150 can all have the same fixed capacitances, or they may havedifferent fixed capacitances. To set the variable capacitance of each ofthe variable capacitive elements 128′, 130′, 132′, 134′, 148, and 150, aparticular combination of the selectable capacitive components from atotal set of permissible combinations is selected in each of thevariable capacitive elements 128′, 130′, 132′, 134′, 148, and 150. Sincethe selectable capacitive components each have fixed capacitances, thevariable capacitance of each of the variable capacitive elements 128′,130′, 132′, 134′, 148, and 150 varies in a discrete manner depending onthe particular combination of the selectable capacitive components thathas been selected.

The hybrid control output 111 thus includes hybrid word outputs 111-1′,111-2′, 111-3′, 111-4′, 111-5′, 111-6′ for the variable capacitiveelements 128′, 130′, 132′, 134′, 148, and 150. Each one of the hybridword outputs 111-1′, 111-2′, 111-3′, 111-4′, 111-5′, 111-6′ may includeone or more bit signals so that each one of the hybrid word outputs111-1′, 111-2′, 111-3′, 111-4′, 111-5′, 111-6′ represents a decodedword. The decoded word indicates the particular combination ofselectable capacitive components that are to be selected in acorresponding one of the variable capacitive elements 128′, 130′, 132′,134′, 148, and 150. Thus, the decoded word represents a particulardiscrete capacitance value for the variable capacitance of thecorresponding one of the variable capacitive elements 128′, 130′, 132′,134′, 148, and 150. Each of the variable capacitive elements 128′, 130′,132′, 134′, 148, and 150 is configured to set the variable capacitancein accordance with a corresponding one of the hybrid word outputs111-1′, 111-2′, 111-3′, 111-4′, 111-5′, 111-6′.

For example, the variable capacitive element 128′ is configured to setthe first variable capacitance in accordance to a decoded wordrepresented by the hybrid word output 111-1′. The decoded wordrepresents a particular combination of the selectable capacitivecomponents in the variable capacitive element 128′. The variablecapacitive element 128′ is configured to select the particularcombination of the selectable capacitive components in the variablecapacitive element 128′ in response to the hybrid word output 111-1′.The variable capacitive element 130′ is configured to set the secondvariable capacitance in accordance to a decoded word represented by thehybrid word output 111-2′. The decoded word represents a particularcombination of the selectable capacitive components in the variablecapacitive element 130′. The variable capacitive element 130′ isconfigured to select the particular combination of the selectablecapacitive components in the variable capacitive element 130′ inresponse to the hybrid word output 111-2′. The variable capacitiveelement 132′ is configured to set the third variable capacitance inaccordance to a decoded word represented by the hybrid word output111-3′. The decoded word represents a particular combination of theselectable capacitive components in the variable capacitive element132′. The variable capacitive element 132′ is configured to select theparticular combination of the selectable capacitive components in thevariable capacitive element 132′ in response to the hybrid word output111-3′. The variable capacitive element 134′ is configured to set thefourth variable capacitance in accordance to a decoded word representedby the hybrid word output 111-4′. The decoded word represents aparticular combination of the selectable capacitive components in thevariable capacitive element 134′. The variable capacitive element 134′is configured to select the particular combination of the selectablecapacitive components in the variable capacitive element 134′ inresponse to the hybrid word output 111-4′. The variable capacitiveelement 148 is configured to set the fifth variable capacitance inaccordance to a decoded word represented by the hybrid word output111-5′. The decoded word represents a particular combination of theselectable capacitive components in the variable capacitive element 148.The variable capacitive element 148 is configured to select theparticular combination of the selectable capacitive components in thevariable capacitive element 148 in response to the hybrid word output111-5′. The variable capacitive element 150 is configured to set thesixth variable capacitance in accordance to a decoded word representedby the hybrid word output 111-6′. The decoded word represents aparticular combination of the selectable capacitive components in thevariable capacitive element 150. The variable capacitive element 150 isconfigured to select the particular combination of the selectablecapacitive components in the variable capacitive element 150 in responseto the hybrid word output 111-6′.

Thus, the first variable capacitance, the second variable capacitance,the third variable capacitance, the fourth variable capacitance, thefifth variable capacitance, and the sixth variable capacitance are eachconfigured to be adjusted in accordance with the respective decoded wordrepresented by the corresponding one of the respective hybrid wordoutputs 111-1′, 111-2′, 111-3′, 111-4′, 111-5′, 111-6′. In this manner,the tunable hybrid coupler 114′ can be tuned to increase impedancematching between input and output impedances at the ports 116, 118,120,122 within the different RF communication bands.

FIG. 9B illustrates one embodiment of the tunable hybrid coupler 114′formed on the semiconductor substrate 136. The first port 116, thesecond port 118, the third port 120, the fourth port 122, the firstinductive element 124, and the second inductive element 126 are eachformed in the same manner described above with respect to the tunablehybrid coupler 114 shown in FIG. 8B. However, unlike the tunable hybridcoupler 114 illustrated in FIG. 8B, each of the variable capacitiveelements 128′, 130′, 132′, and 134′ in the tunable hybrid coupler 114′illustrated in FIG. 9B is formed as a PCA in the semiconductor substrate136. Additionally, the tunable hybrid coupler 114′ has the variablecapacitive elements 148, 150, each of which is also formed as a PCA inthe semiconductor substrate 136. The variable capacitive element 148 isconnected in parallel with the portion 152 of the first inductiveelement 124. More specifically, the variable capacitive element 148 ofFIG. 9B is connected from the first port 116 to the intermediary node154 of the first inductive element 124. Additionally, the variablecapacitive element 150 is connected in parallel with the portion 156 ofthe second inductive element 126. More specifically, the variablecapacitive element 150 of FIG. 9A is connected from the second port 118to the intermediary node 158 of the second inductive element 126. Duringoperation, adjustments can be made to the variable capacitances of thevariable capacitive elements 128′, 130′, 132′, 134′, 148, 150 so as totune the tunable hybrid coupler 114′ to a selected one of the differentRF communication bands.

FIGS. 10A-10E illustrate embodiments of RF filters 160, 162, 164, 166,168, respectively. Each of the RF filters 160, 162, 164, 166, 168, is ashunt coupled RF filter. Accordingly, the first RF filter 106 in the RFfilter circuit 22B of FIGS. 6A and 6B can be provided as any one of theRF filters 160, 162, 164, 166, 168, shown in FIGS. 10A-10E. Similarly,the second RF filter 108 in the RF filter circuit 22B of FIGS. 6A and 6Bcan be provided as any one of the RF filters 160, 162, 164, 166, 168shown in FIGS. 10A-10E.

Referring now to FIG. 10A, as the first RF filter 106 (see FIG. 6A), theRF filter 160 is tunable to pass the first RF QHRS 56 (see FIG. 6A) tothe second hybrid coupler 20′ (see FIG. 6A), or, as the second RF filter108 (see FIG. 6A), the RF filter 160 is tunable to pass the second RFQHRS 58 (see FIG. 6A) to the second hybrid coupler 20′ (see FIG. 6A).Also, as the second RF filter 108 (see FIG. 6B) the RF filter 160 istunable to reflect the first RF QHTS 60 (see FIG. 6B) back to the firsthybrid coupler 18′ (see FIG. 6B), or, as the first RF filter 106 (seeFIG. 6B), the RF filter 160 is tunable to reflect the second RF QHTS 62(see FIG. 6B) back to the first hybrid coupler 18′ (see FIG. 6B). The RFfilter 160 includes a series LC resonator 170 and a variable capacitiveelement 172 that are coupled in shunt between the first hybrid coupler18′ and the second hybrid coupler 20′. The series LC resonator 170 isthus connected in parallel with respect to the variable capacitiveelement 172. The variable capacitive element 172 is operable to providea variable capacitance. Thus, in this embodiment, the variablecapacitive element 172 is coupled to present a variable capacitiveimpedance from a line 173 to ground. The series LC resonator 170includes an inductive element 174 and a variable capacitive element 176coupled in series with respect to one another. The variable capacitiveelement 176 also has a variable capacitance. The series LC resonator 170is therefore tunable by adjusting the variable capacitance of thevariable capacitive element 176. Thus, in this embodiment, the series LCresonator 170 is coupled to present a variable reactive impedance fromthe line 173 to ground.

The RF filter 160 has a frequency response that defines a stopband(i.e., a notch). The series LC resonator 170 operates approximately as ashort circuit at a series resonant frequency of the series LC resonator170. Accordingly, the stopband of the RF filter 160 is transposed,shifting the series resonant frequency, which is done in this embodimentby adjusting the variable capacitance of the variable capacitive element176. To do this, the variable capacitance of the variable capacitiveelement 176 is set in response to the transmission tuning control output112. The transmission tuning control output 112 either represents adecoded word or has a signal level that indicates the variablecapacitance value for setting the variable capacitance of the variablecapacitive element 176. In this manner, the series resonant frequency isset at approximately the transmission frequency of the RF transmissioninput signal 26 (see FIGS. 6A-6B), and the stopband is shifted withinthe RF transmission band of the RF transmission input signal 26 (seeFIGS. 6A-6B).

The variable capacitive element 172 and the series LC resonator 170 forma parallel LC resonator. The RF filter 160 thus operates approximatelyas an open circuit at a parallel resonant frequency of the parallel LCresonator. The frequency response of the RF filter 160 thus also definesa passband. The parallel resonant frequency is equal to approximatelythe series resonant frequency plus a frequency offset. The frequencyoffset is determined in accordance to the variable capacitance of thevariable capacitive element 172. Accordingly, the frequency offset ofthe RF filter 160 from the series resonant frequency can be transposedby adjusting the variable capacitance of the variable capacitive element176. To do this, the variable capacitance of the variable capacitiveelement 176 is set in response to the receive control output 110. Thereceive control output 110 either represents a decoded word or has asignal level that indicates the variable capacitance value for settingthe variable capacitance of the variable capacitive element 172. In thismanner, the parallel resonant frequency is set at approximately thereceive frequency of the RF receive input signal 14 (see FIGS. 6A-6B),and the passband is shifted within the RF receive band of the RF receiveinput signal 14 (see FIGS. 6A-6B). The RF filter 160 demonstrates goodperformance when operating in 3G and 4G RF communication bands. However,RF communication bands with smaller duplex offsets between the RFtransmission band and the RF communication band, such as Long TermEvolution (LTE) RF communication bands, require a higher quality factorperformance.

FIG. 10B illustrates an embodiment of the RF filter 162. As the first RFfilter 106 (see FIG. 6A), the RF filter 162 is tunable to pass the firstRF QHRS 56 (see FIG. 6A) to the second hybrid coupler 20′ (see FIG. 6A),or, as the second RF filter 108 (see FIG. 6A), the RF filter 162 istunable to pass the second RF QHRS 58 (see FIG. 6A) to the second hybridcoupler 20′ (see FIG. 6A). Also, as the second RF filter 108 (see FIG.6B), the RF filter 162 is tunable to reflect the first RF QHTS 60 (seeFIG. 6B) back to the first hybrid coupler 18′ (see FIG. 6B), or, as thefirst RF filter 106 (see FIG. 6B), the RF filter 162 is tunable toreflect the second RF QHTS 62 (see FIG. 6B) back to the first hybridcoupler 18′ (see FIG. 6B). In this embodiment, the RF filter 162 has aplurality of SAW resonators (referred to generically as element 178, andspecifically as elements 178A-178F). The SAW resonators 178 are formedon a piezo-electric substrate 180. These SAW resonators 178 areselectable by a set of single-pole multiple-throw switches (referred togenerically as element 182, and specifically as elements 182A and 182B).Alternatively, the RF filter 162 (and the RF filters 164, 166, and 168)may only include either the single-pole multiple-throw switch 182A (andnot the single-pole multiple-throw switch 182B) or the single-polemultiple-throw switch 182B (and not the single-pole multiple-throwswitch 182A). In some embodiments, providing only the single-polemultiple-throw switch 182A (and not the single-pole multiple-throwswitch 182B) or the single-pole multiple-throw switch 182B (and not thesingle-pole multiple-throw switch 182A) has provided for a betterquality factor in comparison to using both of the single-polemultiple-throw switches 182A, 182B in series.

Each of the SAW resonators 178 in FIG. 10B corresponds to a different RFcommunication band. Essentially, each of the SAW resonators 178 isconfigured to present an inductance in series with a first capacitanceand a second capacitance in parallel with the inductance in series withthe first capacitance. Each of the SAW resonators 178 thus operatesapproximately as a short circuit at a series resonant frequency of theinductance in series with the first capacitance. Additionally, each ofthe SAW resonators 178 thus operates approximately as a open circuit ata parallel resonant frequency of the second capacitance in parallel withthe inductance in series with the first capacitance. The inductance, thefirst capacitance, and the second capacitance of each of the SAWresonators 178 have different values. As such, a stopband and a passbanddefined by a frequency response of the RF filter 162 are tuneddiscretely by selecting a different one of the SAW resonators 178. TheRF filter 162 shown in FIG. 10B has SAW resonators 178 for six differentRF communication bands. Optionally, each of the SAW resonators 178 isconnected in series with an inductive element (referred to genericallyas elements 184, and specifically as elements 184A-184F). Each of theinductive elements 184 has a different inductance.

The SAW resonator 178A is selectable to set the stopband of the RFfilter 162 within a first RF transmission band of a first RFcommunication band and to set the passband within a first RF receiveband of the first RF communication band. The inductance of the inductiveelement 184A provides a shift in the series resonant frequency and theparallel resonant frequency so as to more accurately set the stopbandand the passband within the first RF communication band. The SAWresonator 178B is selectable to set the stopband of the RF filter 162within a second RF transmission band of a second RF communication bandand to set the passband within a second RF receive band of the second RFcommunication band. The inductance of the inductive element 184Bprovides a shift in the series resonant frequency and the parallelresonant frequency so as to more accurately set the stopband and thepassband within the second RF communication band. The SAW resonator 178Cis selectable to set the stopband of the RF filter 162 within a third RFtransmission band of a third RF communication band and to set thepassband within a third RF receive band of the third RF communicationband. The inductance of the inductive element 184C provides a shift inthe series resonant frequency and the parallel resonant frequency so asto more accurately set the stopband and the passband within the third RFcommunication band. The SAW resonator 178D is selectable to set thestopband of the RF filter 162 within a fourth RF transmission band of afourth RF communication band and to set the passband within a fourth RFreceive band of the fourth RF communication band. The inductance of theinductive element 184D provides a shift in the series resonant frequencyand the parallel resonant frequency so as to more accurately set thestopband and the passband within the fourth RF communication band. TheSAW resonator 178E is selectable to set the stopband of the RF filter162 within a fifth RF transmission band of a fifth RF communication bandand to set the passband within a fifth RF receive band of the fifth RFcommunication band. The inductance of the inductive element 184Eprovides a shift in the series resonant frequency and the parallelresonant frequency so as to more accurately set the stopband and thepassband within the fifth RF communication band. The SAW resonator 178Fis selectable to set the stopband of the RF filter 162 within a sixth RFtransmission band of a sixth RF communication band and to set thepassband within a sixth RF receive band of the sixth RF communicationband. The inductance of the inductive element 184E provides a shift inthe series resonant frequency and the parallel resonant frequency so asto more accurately set the stopband and the passband within the sixth RFcommunication band.

In this embodiment, the series LC resonator 170 described above withrespect to FIG. 10A is connected in series with the single-polemultiple-throw switch 182A. Thus, the series LC resonator 170 isconfigured to be connected in series with each of the SAW resonators 178when a particular one of the SAW resonators 178 is selected. Asmentioned above, the stopband and the passband defined by the frequencyresponse of the RF filter 162 are tuned discretely by selecting one ofthe different SAW resonators 178. The series LC resonator 170 shown inFIG. 10B is tunable so as to further shift the series resonant frequencyof the selected one of the SAW resonators 178. In this manner, theseries LC resonator 170 is configured to provide additional and moregranular tuning of the series resonant frequency. The transmissiontuning control output 112 either represents a decoded word or has asignal level that indicates the variable capacitance value for settingthe variable capacitance of the variable capacitive element 176. Thevariable capacitive element 172 described above with respect to FIG. 10Aalso is connected in shunt with the line 173 in FIG. 10B. In thismanner, the second variable capacitor is tunable so as to adjust thefrequency offset of the parallel resonant frequency from the seriesresonant frequency. The receive control output 110 either represents adecoded word or has a signal level that indicates the variablecapacitance value for setting the variable capacitance of the variablecapacitive element 172.

To select each of the SAW resonators 178, the transmission tuningcontrol output 112 may include a decoded word to operate the single-polemultiple-throw switch 182A and may include a decoded word to operate thesingle-pole multiple-throw switch 182B. The decoded words indicate whichof the SAW resonators 178 is to be selected. The single-polemultiple-throw switch 182A and the single-pole multiple-throw switch182B are configured to select the appropriate one of the SAW resonators178 in response to the decoded words in the transmission tuning controloutput 112. The RF filter 162 with the SAW resonators 178 provides verygood system performance with LTE communication bands because the SAWresonators 178 have a higher quality factor performance. However, the RFfilter 160 shown in FIG. 10A may provide better performance with respectto 3G/4G RF communication bands with larger duplex offsets than the RFfilter 162 shown in FIG. 10B.

FIG. 10C illustrates the RF filter 164. As the first RF filter 106 (seeFIG. 6A), the RF filter 164 is tunable to pass the first RF QHRS 56 (seeFIG. 6A) to the second hybrid coupler 20′ (see FIG. 6A), or, as thesecond RF filter 108 (see FIG. 6A) the RF filter 164 is tunable to passthe second RF QHRS 58 (see FIG. 6A) to the second hybrid coupler 20′(see FIG. 6A). Also, as the second RF filter 108 (see FIG. 6B), the RFfilter 164 is tunable to reflect the first RF QHTS 60 (see FIG. 6B) backto the first hybrid coupler 18′ (see FIG. 6B), or, as the first RFfilter 106 (see FIG. 6B), the RF filter 164 is tunable to reflect thesecond RF QHTS 62 (see FIG. 6B) back to the first hybrid coupler 18′(see FIG. 6B). The RF filter 164 is an amalgamation of the RF filter 160shown in FIG. 10A and the RF filter 162 shown in FIG. 10B. Inparticular, the RF filter 164 includes a switchable ground path 186connected in shunt between the series LC resonator 170 and thesingle-pole multiple-throw switch 182A. The switchable ground path 186includes a switch 188. When the switch 188 is in a closed state, the SAWresonators 178 are bypassed and the RF filter 164 operates in the samemanner as the RF filter 160 described above with respect to FIG. 10A.However, when the switch 188 is in an open state, the RF filter 164operates in the same manner as the RF filter 162 described above withrespect to FIG. 10B.

FIG. 10D illustrates the RF filter 166. As the first RF filter 106 (seeFIG. 6A), the RF filter 166 is tunable to pass the first RF QHRS 56 (seeFIG. 6A) to the second hybrid coupler 20′ (see FIG. 6A), or, as thesecond RF filter 108 (see FIG. 6A), the RF filter 166 is tunable to passthe second RF QHRS 58 (see FIG. 6A) to the second hybrid coupler 20′(see FIG. 6A). Also, as the second RF filter 108 (see FIG. 6B), the RFfilter 166 is tunable to reflect the first RF QHTS 60 (see FIG. 6B) backto the first hybrid coupler 18′ (see FIG. 6B), or, as the first RFfilter 106 (see FIG. 6B) the RF filter 166 is tunable to reflect thesecond RF QHTS 62 (see FIG. 6B) back to the first hybrid coupler 18′(see FIG. 6B). The RF filter 166 is similar to the RF filter 162described above with respect to FIG. 10B except that the RF filter 166of FIG. 10D does not have the variable capacitive element 172 connectedin shunt with respect to the line 173. Instead, the RF filter 166 has aparallel LC resonator 190 connected in shunt between the series LCresonator 170 and the single-pole multiple-throw switch 182A. As such,the parallel LC resonator 190 is configured to be connected in parallelwith each of the SAW resonators 178 when one of the SAW resonators 178is selected. In this manner, the parallel LC resonator 190 is tunable soas to adjust the frequency offset of the parallel resonant frequency ofthe selected one of the SAW resonators 178 from the series resonantfrequency of the selected one of the SAW resonators 178.

The parallel LC resonator 190 shown in FIG. 10D includes an inductiveelement 192 and a variable capacitive element 194. The inductive element192 and the variable capacitive element 194 are connected in parallelwith respect to one another. The variable capacitive element 194 has avariable capacitance and the inductive element 192 has a fixedinductance. To tune the parallel LC resonator 190, the receive controloutput 110 either represents a decoded word or has a signal level thatindicates the variable capacitance value for setting the variablecapacitance of the variable capacitive element 194.

FIG. 10E illustrates the RF filter 168. As the first RF filter 106 (seeFIG. 6A), the RF filter 168 is tunable to pass the first RF QHRS 56 (seeFIG. 6A) to the second hybrid coupler 20′ (see FIG. 6A), or, as thesecond RF filter 108 (see FIG. 6A), the RF filter 168 is tunable to passthe second RF QHRS 58 (see FIG. 6A) to the second hybrid coupler 20′(see FIG. 6A). Also, as the second RF filter 108 (see FIG. 6B), the RFfilter 168 is tunable to reflect the first RF QHTS 60 (see FIG. 6B) backto the first hybrid coupler 18′ (see FIG. 6B), or, as the first RFfilter 106 (see FIG. 6B), the RF filter 168 is tunable to reflect thesecond RF QHTS 62 (see FIG. 6B) back to the first hybrid coupler 18′(see FIG. 6B). The RF filter 168 is similar to the RF filter 166described above with respect to FIG. 10D except that instead of havingthe parallel LC resonator 190 of FIG. 10D, the RF filter 168 of FIG. 10Ehas a series LC resonator 196. More specifically, the series LCresonator 196 is connected in shunt between the series LC resonator 170and the single-pole multiple-throw switch 182A. As such, the series LCresonator 196 is configured to be connected in parallel with each of theSAW resonators 178 when a selected one of the SAW resonators 178 isselected. In this manner, the series LC resonator 196 is tunable so asto adjust the frequency offset of the parallel resonant frequency of theselected one of the SAW resonators 178 from the series resonantfrequency of the selected one of the SAW resonators 178.

The series LC resonator 196 shown in FIG. 10E includes an inductiveelement 198 and a variable capacitive element 200. The inductive element198 and the variable capacitive element 200 are connected in series withrespect to one another. The variable capacitive element 200 has avariable capacitance and the inductive element 198 has a fixedinductance. To tune the parallel LC resonator 200, the receive controloutput 110 either represents a decoded word or has a signal level thatindicates the variable capacitance value for setting the variablecapacitance of the variable capacitive element 200.

Referring now to FIG. 11A, FIG. 11A illustrates an exemplary frequencyresponse 202 of any one of the RF filters 160, 162, 164, 166, 168 shownin FIGS. 10A-10E. The frequency response 202 defines a stopband 204 as aresult of the series resonant frequency (SRF). A minima 206 of thestopband 204 is set at the series resonant frequency (SRF). Changing theseries resonant frequency (SRF) thereby shifts the stopband 204. Thefrequency response 202 for each of the RF filters 160, 162, 164, 166,168 also defines a passband 208 where a maxima 210 of the passband 208is set at the parallel resonant frequency (PRF). However, the parallelresonant frequency (PRF) is a function of the series resonant frequency(SRF). As previously discussed, the parallel resonant frequency (PRF) isthe series resonant frequency (SRF) plus the frequency offset (FO).

Referring now to FIGS. 11A and 11B, FIG. 11B illustrates an RFcommunication channel (RFCC) having an RF receive band (RFRB) and an RFtransmission band (RFTB) separated by a duplex offset (DO). The RFreceive band (RFRB) is has a higher frequency range than the RFtransmission band (RFTB). By tuning the series resonant frequency (SRF)to be at a frequency within the RF transmission band (RFTB), the minima206 is shifted and the stopband 204 is placed within the RF transmissionband (RFTB). The frequency offset (FO) can then be adjusted to set thefrequency offset (FO) to the duplex offset (DO) of an RF communicationchannel (RFCC). In this manner, the maxima 210 is shifted and thepassband 208 is placed within the RF receive band (RFRB) of the RFcommunication channel (RFCC). However, the frequency offset (FO)provided by the RF filters 160, 162, 164, 166, 168 shown in FIGS.10A-10E cannot be negative. Thus, the RF filters 160, 162, 164, 166, 168are configured to operate with the RF communication channels, like theRF communication channel (RFCC), where the RF receive band (RFRB) hasthe higher frequency range in comparison to the RF transmission band(RFTB). Most RF communication channels defined by RF communicationspecifications and/or RF communication standards have the RF receiveband (RFRB) with the higher frequency range. However, a few RFcommunication channels, particularly certain RF communication channelsdefined by LTE RF communication standards, have an RF transmission bandat a higher frequency range than the RF receive band. In this case, theRF filters 160, 162, 164, 166, 168 in and of themselves would not beable to set the passband 208 within the RF receive band at a lowerfrequency range than the RF transmission band. This is because thefrequency offset (FO) has to be positive and thus the passband 208 mustbe set at a higher frequency range than the stopband 204. Since thestopband 204 is tuned within the RF transmission band and the frequencyoffset (FO) has to be positive, the passband 208 cannot be set withinthe RF receive band at a lower frequency range than the RF transmissionband.

FIGS. 12A-12C illustrate exemplary embodiments of a tunable RF duplexer10C, a tunable RF duplexer 10D, and a tunable RF duplexer 10E,respectively. Like the tunable RF duplexer 10B shown in FIGS. 6A and 6B,the tunable RF duplexers 10C-10E include the first RF filter 106 and thesecond RF filter 108. In each of the tunable RF duplexers 10C-10E, thefirst RF filter 106 may be provided as any of the RF filters 160, 162,164, 166, 168 described above with regard to FIGS. 10A-10E. Also, ineach of the tunable RF duplexers 10C-10E, the second RF filter 108 maybe provided as any of the RF filters 160, 162, 164, 166, 168 describedabove with regard to FIGS. 10A-10E. Each of the tunable RF duplexers10C, 10D, and 10E is tunable to set the stopband within the RFtransmission band and to set the passband within the RF receive bandwhenever the RF receive band is at a higher frequency range than the RFtransmission band. However, each of the tunable RF duplexers 10C, 10D,and 10E can also handle RF communication channels where the RF receiveband is at a lower frequency range than the RF transmission band.

Referring now to FIG. 12A, the tunable RF duplexer 10C is like thetunable RF duplexer 10B in FIGS. 6A and 6B, except that the tunable RFduplexer 10C includes an RF filter circuit 22C. The RF filter circuit22C has the same first RF filter 106 and the same second RF filter 108as the RF filter circuit 22B in FIGS. 6A and 6B. However, the RF filtercircuit 22C further includes a switchable bypass path 212, aquarter-wave phase-shifting transmission line element 214, a switchablebypass path 216, and a quarter-wave phase-shifting transmission lineelement 218. The quarter-wave phase-shifting transmission line element214 is connected between a line 173A and the first RF filter 106. Theline 173A connects the third port 80 of the first hybrid coupler 18′ tothe fifth port 84 of the second hybrid coupler 20′. In FIG. 12A, thequarter-wave phase-shifting transmission line element 214 is connectedin series with the first RF filter 106. The switchable bypass path 212is connected from the line 173A to the first RF filter 106. Accordingly,the quarter-wave phase-shifting transmission line element 214 isconnected in parallel with respect to the switchable bypass path 212. Assuch, the switchable bypass path 212 is coupled such that thequarter-wave phase-shifting transmission line element 214 is bypassedwhen the switchable bypass path 212 is closed, and is not bypassed whenthe switchable bypass path 212 is open.

Additionally, the quarter-wave phase-shifting transmission line element218 is connected between a line 173B and the second RF filter 108. Theline 173B connects the fourth port 82 of the first hybrid coupler 18′ tothe sixth port 86 of the second hybrid coupler 20′. In FIG. 12A, thequarter-wave phase-shifting transmission line element 218 is connectedin series with the second RF filter 108. The switchable bypass path 216is connected from the line 173B to the second RF filter 108.Accordingly, the quarter-wave phase-shifting transmission line element218 is connected in parallel with respect to the switchable bypass path216. As such, the switchable bypass path 216 is coupled such that thequarter-wave phase-shifting transmission line element 218 is bypassedwhen the switchable bypass path 216 is closed, and is not bypassed whenthe switchable bypass path 216 is open.

When the switchable bypass path 212 and the switchable bypass path 216are both open, the tunable RF duplexer 10C operates in the same manneras the tunable RF duplexer 10B described above with respect to FIGS. 6Aand 6B. However, when the switchable bypass path 212 and the switchablebypass path 216 are both closed, impedance transformations of thequarter-wave phase-shifting transmission line element 214 and thequarter-wave phase-shifting transmission line element 218 createfrequency responses that can provide duplexing when the RF receive bandis at a lower frequency range than the RF transmission band. Morespecifically, the impedance transformation of the quarter-wavephase-shifting transmission line element 214 presents a frequencyresponse at the line 173A that is a frequency response inversion of thefrequency response provided by the first RF filter 106. In addition, theimpedance transformation of the quarter-wave phase-shifting transmissionline element 214 218 presents a frequency response at the line 173B thatis a frequency response inversion of the frequency response provided bythe second RF filter 108. When the switchable bypass path 212 is closed,the frequency response of the quarter-wave phase-shifting transmissionline element 214 in series with the first RF filter 106 defines astopband and a passband, where the passband is at a lower frequencyrange than the stopband. Similarly, when the switchable bypass path 216is closed, the frequency response of the quarter-wave phase-shiftingtransmission line element 214 218 in series with the second RF filter108 defines a stopband and a passband, where the passband is at a lowerfrequency range than the stopband.

In this embodiment, the series resonant frequency of the first RF filter106 is configured to set the passband when the switchable bypass path212 is closed due to the frequency response inversion. The seriesresonant frequency of the first RF filter 106 may thus be shifted withinthe RF receive band. Additionally, the series resonant frequency of thesecond RF filter 108 is configured to set the passband when theswitchable bypass path 212 is closed due to the frequency responseinversion. The series resonant frequency of the second RF filter 108 maythus be set within the RF receive band. Thus, the series resonantfrequency of the second RF filter 108 may be shifted within the RFreceive band.

The parallel resonant frequency of the first RF filter 106 sets thestopband of the first RF filter 106 when the switchable bypass path 212is closed and the parallel resonant frequency of the second RF filter108 sets the stopband of the second RF filter 108 when the switchablebypass path 216 is closed. Since the parallel resonant frequency isequal to the series resonant frequency plus the frequency offset, thefrequency offset of the first RF filter 106 may be adjusted so that theparallel resonant frequency is shifted within the RF transmission band.Also, since the parallel resonant frequency is equal to the seriesresonant frequency plus the frequency offset, the frequency offset ofthe second RF filter 108 may be adjusted so that the parallel resonantfrequency is shifted within the RF transmission band. In this manner,the stopband of the first RF filter 106 is shifted within the RFtransmission band and the stopband of the second RF filter 108 isshifted within the RF transmission band. Accordingly, the impedancetransformations of the quarter-wave phase-shifting transmission lineelements 214, 218 allow for the tunable RF duplexer 10C to provideduplexing in RF communication bands where the RF receive band is withina lower frequency range with respect to the RF transmission band.

Referring now to FIG. 12B, the tunable RF duplexer 10D is like thetunable RF duplexer 10B in FIGS. 6A and 6B, except that the tunable RFduplexer 10D includes an RF filter circuit 22D. The RF filter circuit22D has the first RF filter 106 and the second RF filter 108, like theRF filter circuit 22B in FIGS. 6A and 6B. However, the RF filter circuit22D further includes a switchable bypass path 220, a third RF filter 222coupled in series between the third port 80 of the first hybrid coupler18′ and the fifth port 84 of the second hybrid coupler 20′, a switchablebypass path 224, a fourth RF filter 226 coupled in series between thefourth port 82 of the first hybrid coupler 18′ and the sixth port 86 ofthe second hybrid coupler 20′, a switch 228, and a switch 230.

The switchable bypass path 220 is configured such that the third RFfilter 222 is bypassed when the switchable bypass path 220 is closed andis configured such that the third RF filter 222 is not bypassed when theswitchable bypass path 220 is open. Thus, the third RF filter 222 doesnot provide filtering when the switchable bypass path 220 is closed, anddoes provide filtering when the switchable bypass path 220 is open. Theswitch 228 is connected between the line 173A and the first RF filter106, and the switch 228 is connected in series with respect to the firstRF filter 106. Thus, the first RF filter 106 does not provide filteringwhen the switch 228 is open and does provide filtering when the switch228 is closed.

Additionally, the switchable bypass path 224 is configured such that thefourth RF filter 226 is bypassed when the switchable bypass path 224 isclosed and is configured such that the fourth RF filter 226 is notbypassed when the switchable bypass path 224 is open. Thus, the fourthRF filter 226 does not provide filtering when the switchable bypass path224 is closed, and does provide filtering when the switchable bypasspath 224 is open. The switch 230 is connected between the line 173B andthe second RF filter 108, and the switch 230 is connected in series withrespect to the second RF filter 108. Thus, the second RF filter 108 doesnot provide filtering when the switch 230 is open and does providefiltering when the switch 230 is closed.

When the switchable bypass path 220, the switchable bypass path 224, theswitch 228, and the switch 230 are all closed, the tunable RF duplexer10D operates in the same manner as the tunable RF duplexer 10B describedabove with respect to FIGS. 6A and 6B. However, when the switchablebypass path 220, the switchable bypass path 224, the switch 228, and theswitch 230 are all open, the third RF filter 222 and the fourth RFfilter 226 provide filtering, instead of the first RF filter 106 and thesecond RF filter 108. The third RF filter 222 has a frequency responsethat defines a passband and a stopband. However, unlike the first RFfilter 106 and the second RF filter 108, the passband of the third RFfilter 222 is at a lower frequency range than the stopband of the thirdRF filter 222. Since the RF receive band is at a lower frequency rangethan the RF transmission band, the third RF filter 222 is tunable so asto set the stopband within the RF transmission band and so as to set thepassband within the RF receive band. The fourth RF filter 226 also has afrequency response that defines a passband and a stopband. However,unlike the first RF filter 106 and the second RF filter 108, thepassband of the fourth RF filter 226 is at a lower frequency range thanthe stopband of the fourth RF filter 226. Since the RF receive band isat the lower frequency range in comparison to the RF transmission band,the fourth RF filter 226 is tunable so as to set the stopband within theRF transmission band and so as to set the passband within the RF receiveband. When the switchable bypass path 220, the switchable bypass path224, the switch 228, and the switch 230 are all open, the tunable RFduplexer 10D is allowed to handle RF communication channels with RFreceive bands at lower frequency ranges than RF transmission bands.

Referring now to FIG. 12C, the tunable RF duplexer 10E is like thetunable RF duplexer 10B in FIGS. 6A and 6B, except that the tunable RFduplexer 10D includes a single-pole multiple-throw switch 232, and asingle-pole multiple-throw switch 234. The single-pole multiple-throwswitch 232 is coupled to the antenna 12. The single-pole multiple-throwswitch 232 is switchable between a first switch state and a secondswitch state. In the first switch state, the single-pole multiple-throwswitch 232 couples the antenna 12 to the first port 76 of the firsthybrid coupler 18′ and does not couple the antenna 12 to the seventhport 88 of the second hybrid coupler 20′. On the other hand, in thesecond switch state, the single-pole multiple-throw switch 232 couplesthe antenna 12 to the seventh port 88 of the second hybrid coupler 20′and does not couple the antenna 12 to the first port 76 of the firsthybrid coupler 18′.

With regard to the single-pole multiple-throw switch 234, thesingle-pole multiple-throw switch 234 is coupled to the impedance load100. The single-pole multiple-throw switch 234 is also switchablebetween the first switch state and the second switch state. In the firstswitch state, the single-pole multiple-throw switch 234 couples theimpedance load 100 to the seventh port 88 of the second hybrid coupler20′ and does not couple the impedance load 100 to the first port 76 ofthe first hybrid coupler 18′. On the other hand, in the second switchstate, the single-pole multiple-throw switch 234 couples the impedanceload 100 to the first port 76 of the first hybrid coupler 18′ and doesnot couple the impedance load 100 to the seventh port 88 of the secondhybrid coupler 20′.

When the single-pole multiple-throw switch 232 and the single-polemultiple-throw switch 234 are both in the first switch state, thetunable RF duplexer 10E operates in the same manner as the tunable RFduplexer 10B described above with respect to FIGS. 6A and 6B. However,when the single-pole multiple-throw switch 232 and the single-polemultiple-throw switch 234 are both in the second switch state, thetransmission signal flow and the receive signal flow are different andallow the tunable RF duplexer 10E to provide duplexing when the RFreceive band is in a lower frequency range than the RF transmissionband. Since the RF receive band is at the lower frequency range incomparison to the RF transmission band when both the single-polemultiple-throw switches 232, 234 are in the second switch state, thetuning circuit 24B is configured to adjust the series resonant frequencyof the first RF filter 106 into the RF receive band and to adjust thefrequency offset so that the parallel resonant frequency is within theRF transmission band. Accordingly, the tuning circuit 24B is configuredto tune the stopband of the first RF filter 106 within the RF receiveband and to tune the passband of the first RF filter 106 within the RFtransmission band.

Also, when both the single-pole multiple-throw switches 232, 234 are inthe second switch state, the tuning circuit 24B is configured to adjustthe series resonant frequency of the second RF filter 108 into the RFreceive band and to adjust the frequency offset so that the parallelresonant frequency is within the RF transmission band. Accordingly, thetuning circuit 24B is configured to tune the stopband of the second RFfilter 108 within the RF receive band and to tune the passband of thesecond RF filter 108 within the RF transmission band. As such, thetuning circuit 24B is configured to tune the stopband of the RF filtercircuit 22B within the RF receive band and to tune the passband of theRF filter circuit 22B within the RF transmission band when both thesingle-pole multiple-throw switches 232, 234 are in the second switchstate.

FIG. 13A illustrates the transmission signal flow of the tunable RFduplexer 10E when the single-pole multiple-throw switch 232 and thesingle-pole multiple-throw switch 234 are both in the second switchstate. When both the single-pole multiple-throw switches are in thesecond switch state, the tuning circuit 24B is configured to tune thefrequency response of the RF filter circuit 22B so that the stopband iswithin the RF receive band and the passband is within the RFtransmission band. When the single-pole multiple-throw switch 232 is inthe second switch state, the RF transmission output signal 16 is outputfrom the seventh port 88 to the antenna 12. The first hybrid coupler 18′is configured to receive the RF transmission input signal 26 at thesecond port 78. Since the single-pole multiple-throw switch 234 is inthe second switch state, the first port 76 is coupled to the impedanceload 100. The first port 76 of the first hybrid coupler 18′ thusisolated from the second port 78. This means that the second port 78 issubstantially unresponsive to signals incident at the first port 76, andthe first port 76 is substantially unresponsive to signals incident atthe second port 78 when the single-pole multiple-throw switch 234 is inthe second switch state. As a result, the first port 76 is substantiallyunresponsive to the RF transmission input signal 26 incident at thesecond port 78.

The first hybrid coupler 18′ is operable to split the RF transmissioninput signal 26 into the first RF QHTS 60 and the second RF QHTS 62. Inthis manner, the first RF QHTS 60 and the second RF QHTS 62 haveapproximately the same power ratio with respect to the RF transmissioninput signal 26, but have a quadrature phase difference of approximately90 degrees or π/2 radians. In the embodiment shown in FIG. 13A, thesecond port 78 is phase-aligned with the fourth port 82, while the thirdport 80 has a quadrature phase shift with respect to the second port 78.Thus, the first RF QHTS 60 is approximately phase-aligned with the RFtransmission input signal 26, but there is a quadrature phase differencebetween the RF transmission input signal 26 and the second RF QHTS 62.

Note that in alternative embodiments, this may or may not be the case.For example, there may be a phase shift between the second port 78 andthe fourth port 82. The phase shift between the second port 78 and thethird port 80 may then be equal to this phase shift plus 90 degrees orπ/2 radians. Accordingly, so long as the phase difference between thefirst RF QHTS 60 and the second RF QHTS 62 is about 90 degrees or π/2radians, phase alignment between the fourth port 82 and the second port78, and between the third port 80 and the second port 78, can vary.

The first RF QHTS 60 is output at the fourth port 82 to the RF filtercircuit 22B. Additionally, the second RF QHTS 62 is output at the thirdport 80 to the RF filter circuit 22B. In this embodiment, the RF filtercircuit 22B has the second RF filter 108 and the first RF filter 106.The second RF filter 108 is coupled to the fourth port 82 so as toreceive the first RF QHTS 60 from the first hybrid coupler 18′. Thefirst RF filter 106 is coupled to the third port 80 so as to receive thesecond RF QHTS 62 from the first hybrid coupler 18′. The second RFfilter 108 and the first RF filter 106 each have a frequency response.The frequency response of the RF filter circuit 22B is thus determinedin accordance with the combined effect of the independent frequencyresponses provided by the second RF filter 108 and the first RF filter106. However, in this embodiment, the second RF filter 108 and the firstRF filter 106 are identical to one another. Furthermore, each is tunedin accordance with a transmission tuning control output 112 from thetuning circuit 24B. As such, the overall frequency response of the RFfilter circuit 22B is the same as the independent frequency responsesprovided by the second RF filter 108 and the first RF filter 106.Alternatively, in other embodiments, the second RF filter 108 and thefirst RF filter 106 may be different and/or may be tuned independentlyby the tuning circuit 24B. As such, the different independent frequencyresponses from the second RF filter 108 and the first RF filter 106 maycombine to determine the overall frequency response of the RF filtercircuit 22B.

Referring again to FIG. 13A, the tuning circuit 24B is configured totune the frequency response of the RF filter circuit 22B so that thepassband includes the RF transmission band when both the single-polemultiple-throw switches 232, 234 are in the second switch state. Thetuning circuit 24B thus shifts the passband of the RF filter circuit 22Bto include the RF transmission band. In this manner, the RF filtercircuit 22B is operable to pass the first RF QHTS 60 and the second RFQHTS 62 to the second hybrid coupler 20′. The manner of tuning thefrequency response may depend on the topology of the RF filter circuit22B. For example, the second RF filter 108 and the first RF filter 106shown in FIG. 13A are each passive filters. Accordingly, one or morereactive impedance components (inductive, capacitive, or both) in eachof the first and second RF filters 106, 108 may have a variable reactiveimpedance level. By varying these variable reactive impedance levels,the poles and zeros of the individual frequency responses provided byeach of the first and second RF filters 106, 108 are adjusted. Thisthereby shifts the passband and/or the stopband of the RF filter circuit22B.

As mentioned above, the tuning circuit 24B illustrated in FIG. 13Agenerates the transmission tuning control output 112 and a receivetuning control signal 110. The variable reactive impedance components inboth the second RF filter 108 and the first RF filter 106 are set inaccordance with a signal level of the transmission tuning control output112. In this manner, the stopband is shifted to include the RF receiveband. Similarly, reactive impedance levels of variable reactivecomponents in the second RF filter 108 and the first RF filter 106 areset in accordance to a signal level of the receive tuning control signal110. In this manner, the stopband is set to include the RF receive band.In alternative embodiments, the RF filter circuit 22B may include activeRF filters, SAW filters, or any other type of RF filter or combinationof RF filters that are suitable to provide a desired frequency response.As such, the tuning circuit 24B may employ alternative types of tuningtopologies that are different from the particular filtering topologybeing employed by the RF filter circuit 22B.

By placing the passband in the RF transmission band, the RF filtercircuit 22B passes the first RF QHTS 60 and the second RF QHTS 62 to thesecond hybrid coupler 20′. In this particular embodiment, the second RFfilter 108 passes the first RF QHTS 60 to the second hybrid coupler 20′,while the first RF filter 106 passes the second RF QHTS 62 to the secondhybrid coupler 20′. As mentioned above, the stopband (for example, thenotch) is set in the RF receive band.

Referring again to FIG. 13A, the second hybrid coupler 20′ receives thefirst RF QHTS 60 from the second RF filter 108 at the sixth port 86. Thesecond RF QHTS 62 is received by the second hybrid coupler 20′ from thefirst RF filter 106 at the fifth port 84. As discussed above, the firstRF QHTS 60 and the second RF QHTS 62 have a quadrature phase differenceof about 90 degrees or π/2 radians. Thus, for example, if the first RFQHTS 60 has a phase of zero degrees, the second RF QHTS 62 would have aphase of approximately 90 degrees (or π/2 radians). From the sixth port86 to the eighth port 90, the second hybrid coupler 20′ provides nophase shift. Alternatively, the second hybrid coupler 20′ may beconfigured to provide a phase shift of Δ from the sixth port 86 to theeighth port 90.

The second hybrid coupler 20′ shown in FIG. 13A is configured to providea quadrature phase shift from the fifth port 84 to the eighth port 90.In this example, the phase shift is 90 degrees (or π/2 radians), andthus the second RF QHTS 62 has a phase, as seen from the eighth port 90,of 180 degrees (note that the second RF QHTS 62 was received with aphase of 90 degrees in this example, and thus is seen with a phase of180 degrees with another phase shift of 90 degrees). Alternatively, thephase shift from the fifth port 84 to the eighth port 90 may be Δ+90degrees (or π/2 radians). In any case, the phase difference between thefirst RF QHTS 60 and the second RF QHTS 62 as seen from the eighth port90 is about 180 degrees (note that the first RF QHTS 60 was receivedwith a phase of zero (0) degrees and thus is seen with a phase of zero(0) degrees at the eighth port 90). Accordingly, the quadrature phaseshift at the eighth port 90 from the fifth port 84 results indestructive interference between the first RF QHTS 60 and the second RFQHTS 62 at the eighth port 90. As a result, the first RF QHTS 60 and thesecond RF QHTS 62 are substantially cancelled at the eighth port 90. Inthis manner, the eighth port 90 is substantially isolated fromtransmission signal flow.

The second hybrid coupler 20′ is configured to output the RFtransmission output signal 16 from the seventh port 88 in response tothe first RF QHTS 60 being received from the RF filter circuit 22B atthe sixth port 86 and the second RF QHTS 62 being received from the RFfilter circuit 22B at the fifth port 84. In this particular embodiment,the second hybrid coupler 20′ is configured to pass the second RF QHTS62 from the fifth port 84 to the seventh port 88. The second hybridcoupler 20′ provides no phase shift to the second RF QHTS 62 from thefifth port 84 to the seventh port 88. The second RF QHTS 62 is thuspassed with a phase of 90 degrees to the seventh port 88. Alternatively,the second hybrid coupler 20′ may provide a phase shift of Δ to thesecond RF QHTS 62 when passed from the fifth port 84 to the seventh port88. The second hybrid coupler 20′ is configured to pass the first RFQHTS 60 from the sixth port 86 to the seventh port 88. The second hybridcoupler 20′ provides a quadrature phase shift to the first RF QHTS 60 atthe seventh port 88. In this embodiment, the quadrature phase shift is90 degrees or π/2 radians. Alternatively, if a phase shift of Δ wasprovided to the second RF QHTS 62 from the fifth port 84 to the seventhport 88, the quadrature phase shift would be Δ+90 degrees (or π/2radians).

Accordingly, the first RF QHTS 60 is provided substantially as aduplicate of the second RF QHTS 62 at the seventh port 88. This is aresult of the quadrature phase shift provided to the first RF QHTS 60when passed from the sixth port 86 to the seventh port 88 (now, at theseventh port 88, the first RF QHTS 60 is shifted to have a phase of 90degrees, just like the second RF QHTS 62). Referring again to theexample discussed previously, if the first RF QHTS 60 has a phase ofzero (0) degrees at the sixth port 86, then the first RF QHTS 60 has aphase of 90 degrees at the seventh port 88. If the phase of the secondRF QHTS 62 at the fifth port 84 is 90 degrees, then the phase of thesecond RF QHTS 62 is also 90 degrees at the seventh port 88.Accordingly, the first RF QHTS 60 is substantially a duplicate of thesecond RF QHTS 62 at the seventh port 88 because the first RF QHTS 60and the second RF QHTS 62 become phase-aligned at the seventh port 88.As a result, the first RF QHTS 60 and the second RF QHTS 62constructively interfere at the seventh port 88 to output the RFtransmission output signal 16 from the seventh port 88. Note that sincethe first RF QHTS 60 and the second RF QHTS 62 substantially cancel atthe eighth port 90 due to destructive interference, very little or nopower is transferred from the first RF QHTS 60 and the second RF QHTS 62to the eighth port 90. Instead most, if not all, of the power in thefirst RF QHTS 60 and the second RF QHTS 62 is transferred to the seventhport 88 and provided in the RF transmission output signal 16. Theantenna 12 is operably associated with the seventh port 88. In thismanner, the antenna 12 can radiate the RF transmission output signal 16to transmit data and information to external communication systems.

The first hybrid coupler 18′ is thus configured to output the first RFQHTS 60 from the fourth port 82 and the second RF QHTS 62 from the thirdport 80 in response to the RF transmission input signal 26 beingreceived at the second port 78. The RF filter circuit 22B, however,provides isolation to the second port 78 from the second hybrid coupler20′ and, as explained in further detail below, from the receive signalflow. Once the first RF QHTS 60 and the second RF QHTS 62 pass throughand are filtered by the RF filter circuit 22B, the second hybrid coupler20′ is configured to output the RF transmission output signal 16 fromthe seventh port 88 in response to the first RF QHTS 60 being receivedfrom the RF filter circuit 22B at the sixth port 86 and the second RFQHTS 62 being received from the RF filter circuit 22B at the fifth port84.

FIG. 13B illustrates the tunable RF duplexer 10E along with the receivesignal flow. The antenna 12 intercepts the RF receive input signal 14 aselectromagnetic waves in free space. These electromagnetic waves resultin excitations within the antenna 12, thereby converting theelectromagnetic waves into the RF receive input signal 14. The secondhybrid coupler 20′ is configured to receive the RF receive input signal14 at the seventh port 88. The RF receive input signal 14 operates inthe RF receive band within the same RF communication band as the RFtransmission input signal 26 (shown in FIG. 13A). The second hybridcoupler 20′ is operable to split the RF receive input signal 14 into thefirst RF QHRS 56 and the second RF QHRS 58. Since the first RF QHRS 56and the second RF QHRS 58 are quadrature hybrids, the first RF QHRS 56and the second RF QHRS 58 are approximately equal in power, but have aquadrature phase difference of 90 degrees or π/2 radians. The secondhybrid coupler 20′ outputs the first RF QHRS 56 from the fifth port 84and outputs the second RF QHRS 58 from the sixth port 86 in response toreceiving the RF receive input signal 14 at the seventh port 88.

In the embodiment illustrated in FIG. 13B, the first RF QHRS 56 isphase-aligned with the RF receive input signal 14, while the second RFQHRS 58 has a phase difference of about 90 degrees with respect to theRF receive input signal 14. It should be noted that this may or may notbe the case. For example, in alternative embodiments, a phase shift of Δmay be provided between the seventh port 88 and the fifth port 84, andthus, a phase shift of Δ+90 degrees (or π/2 radians) would be providedbetween the seventh port 88 and the sixth port 86.

The RF filter circuit 22B is operable to reflect the first RF QHRS 56and the second RF QHRS 58. As discussed above, the frequency response ofthe RF filter circuit 22B defines the stopband and the RF filter circuit22B is tunable so as to shift the stopband. For example, the stopbandmay be a notch that is shiftable. The tuning circuit 24B is configuredto tune the frequency response of the RF filter circuit 22B so that thesignal bandwidth of the first RF QHRS 56 and the signal bandwidth of thesecond RF QHRS 58 are each within the stopband. For instance, the tuningcircuit 24B may be configured to place the notch within the RF receiveband so that the notch is centered at the RF receive signal frequency.In this embodiment, the tuning circuit 24B generates the receive tuningcontrol signal 110. Variable reactive impedance components in both thesecond RF filter 108 and the first RF filter 106 are responsive to thesignal level of the receive tuning control signal 110 so as to adjustthe variable impedance levels based on the signal level of the receivetuning control signal 110. As a result, the notch defined by theindividual frequency response of the second RF filter 108 is shifted toinclude the signal bandwidth of the second RF QHRS 58. Also, the notchdefined by the individual frequency response of the first RF filter 106is shifted to include the signal bandwidth of the first RF QHRS 56. Inother words, the notches defined by the individual frequency responsesof the second RF filter 108 and the first RF filter 106 are placed inthe RF receive band.

Since the tuning circuit 24B has tuned the frequency response of the RFfilter circuit 22B so that the stopband includes the RF receive signalband, the RF filter circuit 22B blocks the first RF QHRS 56 and thesecond RF QHRS 58. Accordingly, the second port 78 is substantiallyisolated from the receive signal flow. The RF filter circuit 22B thenreflects the first RF QHRS 56 and the second RF QHRS 58 back to thesecond hybrid coupler 20′. In the embodiment illustrated in FIG. 13B,the second RF filter 108 reflects the second RF QHRS 58 back to thesecond hybrid coupler 20′ at the sixth port 86. The first RF filter 106reflects the first RF QHRS 56 back to the second hybrid coupler 20′ atthe fifth port 84.

The second hybrid coupler 20′ is configured to combine the first RF QHRS56 and the second RF QHRS 58 into the RF receive output signal 30. Tocombine the first RF QHRS 56 and the second RF QHRS 58 into the RFreceive output signal 30, the second hybrid coupler 20′ is configured topass the second RF QHRS 58 from the sixth port 86 to the eighth port 90.Additionally, the second hybrid coupler is configured to pass the firstRF QHRS 56 from the fifth port 84 to the eighth port 90. However, thesecond hybrid coupler 20′ provides a quadrature phase shift to the firstRF QHRS 56 at the eighth port 90. Thus, the first RF QHRS 56 is providedsubstantially as a duplicate of the second RF QHRS 58 at the eighth port90. For example, if the phase of the second RF QHRS 58 is 90 degrees atthe sixth port 86, the second RF QHRS 58 has a phase of 90 degrees atthe eighth port 90. As such, the phase of the first RF QHRS 56 at thefifth port 84 is approximately zero (0) degrees. However, due to thequadrature phase shift between the fifth port 84 and the eighth port 90,the first RF QHRS 56 has a phase of about 90 degrees at the eighth port90. Accordingly, the first RF QHRS 56 and the second RF QHRS 58constructively interfere at the eighth port 90 to output the RF receiveoutput signal 30 from the eighth port 90.

Also, note that the second hybrid coupler 20′ is configured such thatthe quadrature phase shift at the seventh port 88 results in destructiveinterference between the first RF QHRS 56 and the second RF QHRS 58.Referring again to the previous example provided, at the seventh port88, the first RF QHRS 56 appears to have a phase of zero (0) degrees,but the second RF QHRS 58 appears to have a phase of 180 degrees. As aresult, the first RF QHRS 56 and the second RF QHRS 58 are substantiallycancelled at the seventh port 88. Consequently, most, if not all, of thepower of the first RF QHRS 56 and the second RF QHRS 58 is transferredto the eighth port 90 and provided in the RF receive output signal 30.The second hybrid coupler 20′ is thus configured to output the RFreceive output signal 30 from the eighth port 90 in response to thefirst RF QHRS 56 being reflected back by the RF filter circuit 22B tothe fifth port 84 and the second RF QHRS 58 being reflected back by theRF filter circuit 22B to the sixth port 86. The configuration of thesecond hybrid coupler 20′ and the RF filter circuit 22B thus providessubstantial isolation between the eighth port 90 and the seventh port88.

As shown in FIGS. 13A and 13B, the seventh port 88 of the second hybridcoupler 20′ is configured to simultaneously receive the RF receive inputsignal 14 and output the RF transmission output signal 16. Accordingly,the RF transmission output signal 16 is output from the seventh port 88while the RF receive input signal 14 is being received at the seventhport 88. The tunable RF duplexer 10E can provide this functionalitybecause the eighth port 90 is substantially isolated from the secondport 78. Accordingly, the transmission signal flow is substantiallyisolated from the receive signal flow.

FIG. 14 illustrates one embodiment of a portion of an RF transceiver 236having a distributed duplex filtering topology. The RF transceiver 236includes another embodiment of a tunable RF duplexer 10F, an RF receivechain 238, and an RF transmission chain 240. In FIG. 14, the RF receivechain 238 and the RF transmission chain 240 are only partiallyillustrated for the sake of clarity. In particular, the RF receive chain238 includes a low noise amplifier (LNA) 242, which is illustrated asbeing within the RF receive chain 238 in FIG. 14. Furthermore, the RFtransmission chain 240 includes an RF power amplifier 244, which isillustrated as being within the RF transmission chain 240 in FIG. 14.The tunable RF duplexer 10F illustrated in FIG. 14 is similar to thetunable RF duplexer 10B described above, except that the tunable RFduplexer 10F of FIG. 14 further includes an antenna impedance tuner 246,a third RF filter 248, a fourth RF filter 250, and a quarter-wavephase-shifting transmission line element 252. The first RF filter 106,the second RF filter 108, the third RF filter 248, and the fourth RFfilter 250 may each be identical to one another. The RF transceiver 236has a distributed duplex filtering topology, which allows each of thefirst RF filter 106, second the RF filter 108, the third RF filter 248,and the fourth RF filter 250 in the tunable RF duplexer 10F shallow, butlow loss.

For example, spurious emissions received at the second port 78,particularly spurious emissions within the RF receive band, can degradeisolation performance. To help reduce spurious emissions, the third RFfilter 248 is connected in shunt with respect to the eighth port 90 ofthe second hybrid coupler 20′. Just like the first RF filter 106 and thesecond RF filter 108, the third RF filter 248 is tunable and has afrequency response that defines a passband and a stopband. The tuningcircuit 24B is configured to tune the third RF filter 248 as a slave ofthe first RF filter 106 and the second RF filter 108. Accordingly, thetuning circuit 24B is operable to tune the third RF filter 248 such thatthe stopband is within the RF transmission band and the passband iswithin the RF receive band. In this manner, spurious emissions withinthe RF transmission band at the eighth port 90 are filtered from the RFreceive output signal 30. The third RF filter 248 is shallow, but isconfigured to reduce out of band emissions enough to operate the LNA 242linearly and reduce intermodulation products that can impact receivesensitivity.

The LNA 242 in the RF receive chain 238 includes a linear amplificationstage 254. The linear amplification stage 254 is operable to receive theRF receive output signal 30 from the eighth port 90 of the second hybridcoupler 20′. The linear amplification stage 254 is configured to amplifythe RF receive output signal 30 in accordance with a linearamplification gain. Note that in FIG. 14, the RF receive output signal30 is received by the linear amplification stage 254 as a single-endedsignal. The linear amplification stage 254 is configured to output theRF receive output signal 30 after amplification as a differential signalfrom differential lines 256A and 256B.

The LNA 242 further includes a tunable high-selectivity bandpass filter258. In this embodiment, the tunable high-selectivity bandpass filter258 has an inductive element with a fix inductance and a variablecapacitive element with a variable capacitance. The inductive elementand the variable capacitive element are coupled between the differentiallines 256A and 256B. The tunable high-selectivity bandpass filter 258 istunable by adjusting the variable capacitance of the variable capacitiveelement. As a result, a passband of the tunable high-selectivitybandpass filter 258 may be set within the RF receive band. Despiteproviding greater selectivity, the tunable high-selectivity bandpassfilter 258 has a higher quality factor than the first RF filter 106, thesecond RF filter 108, the third RF filter 248, and the fourth RF filter250, and thus may be more lossy. However, the tunable high-selectivitybandpass filter 258 is coupled to the differential lines 256A and 256Bfrom the linear amplification stage 254. Accordingly, the lossiness isnot a significant performance issue due to the amplification provided bythe linear amplification stage 254.

The LNA 242 also includes a translational filter 260 coupled to receivethe RF receive output signal 30 that is output from the tunablehigh-selectivity bandpass filter 258. The translational filter 260 isconfigured to translate a frequency response of an intermediatefrequency (IF) filter having a passband at an IF into an RF frequencyresponse with a translated passband within the RF receive band. Thetranslational filter 260 is self-aligned by a local oscillator (LO)signal from an RF local oscillator. The translational filter 260 mayinclude an RF passive mixer along with the IF filter. The RF passivemixer translates the frequency response of the IF filter by mixing theRF receive output signal 30 with the LO signal, which determines an RFcenter frequency. The tunable high-selectivity bandpass filter 258significantly increases the performance of the translational filter 260.In essence, the RF passive mixer can create intermodulation distortiondue to mixing effects with spurious emissions at an LO frequency of theLO signal. Since the RF passive mixer is not infinitely linear,selectivity is added by the tunable high-selectivity bandpass filter258. In this manner, selectivity does not have to be increased withinthe tunable RF duplexer 10F, which may result in unacceptable insertionlosses. After filtering by the translational filter 260, the RF receiveoutput signal 30 is output by the translational filter 260 to a linearamplification stage 261. The linear amplification stage 261 isconfigured to amplify the RF receive output signal 30 in accordance witha linear amplification gain, and then output the RF receive outputsignal 30 to the remainder of the RF receive chain 238 for furtherprocessing.

Small reflections at the seventh port 88 due to an impedance mismatchwith the impedance load 100 can also result in distortion at both thesecond port 78 and the eighth port 90. Additionally, small reflectionsat the first port 76 due to an impedance mismatch with the antenna 12 atthe first port 76 can result in distortion at both the second port 78and the eighth port 90. To help reduce spurious emissions, the fourth RFfilter 250 and the quarter-wave phase-shifting transmission line element252 are connected in shunt with respect to the second port 78 of thefirst hybrid coupler 18′ and in series with respect to one another. Justlike the first RF filter 106 and the second RF filter 108, the fourth RFfilter 250 is tunable and has a frequency response that defines apassband and a stopband. The tuning circuit 24B is configured to tunethe fourth RF filter 250 as a slave of the first RF filter 106 and thesecond RF filter 108. However, the quarter-wave phase-shiftingtransmission line element 252 provides a frequency response inversion toa frequency response of the fourth RF filter 250. As such, the stopbandof the quarter-wave phase-shifting transmission line element 252 incombination with the fourth RF filter 250 is provided within the RFreceive band, while the passband is provided within the RF transmissionband. Accordingly, the tuning circuit 24B is operable to tune the fourthRF filter 250 such that the stopband is within the RF receive band andthe passband is within the RF transmission band. In this manner,spurious emissions within the RF receive band are filtered by the fourthRF filter 250 prior to the RF transmission input signal 26 beingreceived at the second port 78. Furthermore, spurious emissions that arereflected to the second port 78 are also filtered by the quarter-wavephase-shifting transmission line element 252 in combination with thefourth RF filter 250.

The RF transmission input signal 26 is received at the second port 78from the RF power amplifier 244, which is in the RF transmission chain240. The RF power amplifier 244 includes a plurality of RF amplifierstages (referred to generically as element 262, and specifically aselements 262A-262B) coupled in cascode. Accordingly, each of theplurality of RF amplifier stages 262 is operable to provideamplification to the RF transmission input signal 26. In other words, bybeing coupled in cascode, the RF amplifier stages 262 provideamplification to the RF transmission input signal 26 in sequence.

The RF power amplifier 244 shown in FIG. 14 has a power amplificationcircuit 264. The power amplification circuit 264 has an initial RFamplifier stage 262A and a final RF amplifier stage 262B. However, otherembodiments of the RF power amplifier 244 may include any number of RFamplifier stages 262 greater than or equal to two (2). The initial RFamplifier stage 262A is the RF amplifier stage 262 at a beginning of thesequence. The final RF amplifier stage 262B is the RF amplifier stage262 at an end of the sequence. Since at least two RF amplifier stages262 are needed to provide cascoded RF amplifier stages 262, the RF poweramplifier 244 includes at least the initial RF amplifier stage 262A andthe final RF amplifier stage 262B. However, the number of RF amplifierstages 262 may be any integer greater than or equal to two (2). As such,there may be any number of intermediate RF amplifier stages, coupled incascode between the initial RF amplifier stage 262A and the final RFamplifier stage 262B. Since the RF amplifier stages 262 are coupled incascode, the RF amplifier stages 262 provide amplification to the RFtransmission input signal 26 in sequence. Accordingly, the initial RFamplifier stage 262 initially provides amplification to the RFtransmission input signal 26 in accordance with an amplifier gainG_(initial). Once the RF transmission input signal 26 is amplified bythe initial RF amplifier stage 262A in accordance with the amplifiergain G_(initial), the final RF amplifier stage 262B amplifies the RFtransmission input signal 26 in accordance to an amplifier gainG_(final). As such, an aggregated amplifier gain of the plurality of RFamplifier stages 262 as a whole may be described asG_(initial)*G_(final).

The RF power amplifier 244 shown in FIG. 14 is operable to receive theRF transmission input signal 26 from upstream RF circuitry, prior toamplification. As shown in FIG. 14, an RF filter 266A is coupled toreceive the RF transmission input signal 26. The RF filter 266A has afrequency response that defines a stopband, such as a notch. The RFfilter 266A is tunable so as to provide the stopband within the RFreceive band. In this manner, spurious emissions within the RF receiveband from the upstream RF circuitry are reduced. Once the RF filter 266Ahas filtered the RF transmission input signal 26, the initial RFamplifier stage 262A receives the RF transmission input signal 26 andprovides amplification to the RF transmission input signal 26 inaccordance with the amplifier gain G_(initial).

The RF power amplifier 244 also includes an RF filter 266B coupled inseries between the initial RF amplifier stage 262A and the final RFamplifier stage 262B. In other embodiments where the number of RFamplifier stages is greater than two (2), there may be other RF filters,like the RF filter 266B, coupled in series between the RF amplifierstages. The RF filter 266B has a frequency response that defines astopband, such as a notch. The RF filter 266B is tunable so as toprovide the stopband within the RF receive band. In this manner,spurious emissions within the RF receive band from the initial RFamplifier stage 262A are reduced. Thus, by distributing filtering ofspurious emissions within the RF receive band throughout the poweramplification circuit 264, the tunable RF duplexer 10F can have lessselectivity and lower insertion losses. This is important, since thepower amplification circuit 264 may be connected to an RF powerconverter 268. The RF power converter 268 is configured to generate asupply voltage from a power source voltage in order to poweramplification by one or more of the RF amplifier stages 262. The RFpower converter 268 may include a switching circuit in order to generatea pulsed output voltage from the power source voltage and an RF filterthat converts the power source voltage into the supply voltage. Whilesuch systems are power efficient, switching circuits can generate highlevels of spurious emissions. Distributing duplex filtering throughoutthe power amplification circuit 264 allows the tunable RF duplexer 10Fto have lower insertion losses while maintaining adequate isolationbetween the receive and transmission signal flows.

As mentioned above, small reflections at the first port 76 due to animpedance mismatch with the antenna 12 at the first port 76 can resultin distortion at both the second port 78 and the eighth port 90. Thus,the antenna impedance tuner 246 is coupled between the antenna 12 andthe first port 76. Since the RF transmission output signal 16 is in theRF transmission band, the impedance of the antenna 12 may result in aportion of the RF transmission output signal 16 being reflected back tothe first port 76. This not only would degrade the RF transmissionoutput signal 16, but would also degrade the isolation between thereceive signal flow and the eighth port 90. The antenna impedance tuner246 is tunable so as to reduce reflections from the antenna 12 to thefirst port 76. More specifically, an impedance of the antenna 12 may betuned so as to provide impedance matching between the first port 76 andthe antenna 12.

As shown in FIG. 14, a digital calibration circuit 270 is coupled to thetunable RF duplexer 10F. The digital calibration circuit 270 isconfigured to calibrate the tunable RF duplexer 10F. With regard to thefirst hybrid coupler 18′ and the second hybrid coupler 20′, the digitalcalibration circuit 270 performs a hybrid calibration routine for eachof the RF communication bands. During the hybrid calibration routine,the first RF filter 106, the second RF filter 108, the third RF filter248, and the fourth RF filter 250 are set in a neutral state so as to befully transparent within both the RF transmission band and the RFreceive band. An RF tuning signal operating within the RF transmissionband is then injected into the second port 78. Power for the RF tuningsignal is set to be at least 20 dB less than maximum power so that adetector in the digital calibration circuit 270 is not burned. The powerat the seventh port 88 is then measured and the digital calibrationcircuit runs a gradient search for maximum power at the seventh port 88by tuning the first hybrid coupler 18′ and the second hybrid coupler20′. Calibration data for the transmission frequency band is then storedand a correction for the receive frequency band is calculated for boththe first hybrid coupler 18′ and the second hybrid coupler 20′. Inreceive noise critical scenarios, the first hybrid coupler 18′ and thesecond hybrid coupler 20′ are tuned in accordance with the correction.In case of transmission insertion loss or transmission leakage issues,the first hybrid coupler 18′ and the second hybrid coupler 20′ use thecalibration for the RF transmission band.

Once the hybrid calibration routine is completed, a filter calibrationroutine is performed. Initially, the variable capacitance of thevariable capacitive element 176 (see FIGS. 10A-10E) is adjusted untilminimum power is detected at the seventh port 88. Alternatively theforward power between the first port 76 and the antenna impedance tuner246 or between the antenna impedance tuner 246 and the antenna 12 isused to determine a notch frequency for the RF transmission band. Inthat case, the goal is max power at the forward coupled output. Once theappropriate values are determined, the optimal tuning setting is copiedto the RF receive band (λ/4 inverted RF transmission band notch) at thetransmission side and the RF transmission band notch at the receive bandside in front of the LNA 242.

Those skilled in the art will recognize improvements and modificationsto the embodiments of the present disclosure. All such improvements andmodifications are considered within the scope of the concepts disclosedherein and the claims that follow.

What is claimed is:
 1. A radio frequency (RF) transceiver having adistributed duplex filtering topology comprising: a power amplifiercomprising a plurality of RF amplifier stages coupled in cascode and anRF filter coupled between a first one of the plurality of RF amplifierstages and a second one of the plurality of RF amplifier stages,wherein: each of the plurality of RF amplifier stages is configured toamplify an RF transmission input signal that operates within an RFtransmission band of an RF communication band; and the RF filter has afrequency response that defines a stopband and is tunable to provide thestopband within an RF receive band of the RF communication band; and atunable RF duplexer configured to input the RF transmission input signalfrom the power amplifier, generate an RF transmission output signal thatoperates within the RF transmission band in response to the RFtransmission input signal from the power amplifier, and simultaneouslyoutput the RF transmission output signal to an antenna and input an RFreceive input signal that operates within the RF receive band from theantenna.
 2. The RF transceiver of claim 1 further comprising a low noiseamplifier (LNA) wherein: the tunable RF duplexer is further configuredto generate an RF receive output signal that operates within the RFreceive band in response to the RF receive input signal from theantenna; and the LNA comprises a linear amplification stage and atunable high-selectivity bandpass filter; wherein: the linearamplification stage is coupled to receive the RF receive output signalfrom the tunable RF duplexer, and is configured to amplify the RFreceive output signal; and the tunable high-selectivity bandpass filteris tunable to provide a passband within the RF receive band, and iscoupled to receive the RF receive output signal after amplification bythe linear amplification stage.
 3. The RF transceiver of claim 2 whereinthe LNA further comprises a translational filter, the translationalfilter being coupled to receive the RF receive output signal that isoutput from the tunable high-selectivity bandpass filter.
 4. The RFtransceiver of claim 3 wherein the translational filter comprises anintermediate frequency (IF) filter, wherein the translational filter isconfigured to translate a frequency response of the IF filter having apassband at an IF into an RF frequency response with a translatedpassband within the RF receive band.
 5. The RF transceiver of claim 4wherein the translational filter further comprises an RF passive mixerconfigured to mix the RF receive output signal with a local oscillatorsignal.
 6. The RF transceiver of claim 2 wherein the linearamplification stage is configured to receive the RF receive outputsignal as a single-ended signal and output the RF receive output signalas a differential signal.
 7. The RF transceiver of claim 2 wherein thetunable RF duplexer comprises: a first hybrid coupler operable to: splitthe RF transmission input signal into a first RF quadrature hybridtransmission signal (QHTS) and a second RF QHTS; split the RF receiveinput signal into a first RF quadrature hybrid receive signal (QHRS) anda second RF QHRS; and combine the first RF QHTS and the second RF QHTSinto the RF transmission output signal when reflected back to the firsthybrid coupler; a second hybrid coupler configured to combine the firstRF QHRS and the second RF QHRS into the RF receive output signal; and anRF filter circuit that is tunable so as to pass the first RF QHRS andthe second RF QHRS to the second hybrid coupler and to reflect the firstRF QHTS and the second RF QHTS back to the first hybrid coupler.
 8. TheRF transceiver of claim 1 wherein the tunable RF duplexer comprises: afirst hybrid coupler operable to: split the RF transmission input signalinto a first RF quadrature hybrid transmission signal (QHTS) and asecond RF QHTS; split the RF receive input signal into a first RFquadrature hybrid receive signal (QHRS) and a second RF QHRS; andcombine the first RF QHTS and the second RF QHTS into the RFtransmission output signal when reflected back to the first hybridcoupler; a second hybrid coupler configured to combine the first RF QHRSand the second RF QHRS into an RF receive output signal; and an RFfilter circuit that is tunable so as to pass the first RF QHRS and thesecond RF QHRS to the second hybrid coupler and to reflect the first RFQHTS and the second RF QHTS back to the first hybrid coupler.
 9. Thetunable RF duplexer of claim 8 further comprising a tuning circuitwherein: the RF filter circuit has a frequency response that defines apassband; the RF filter circuit is tunable to shift the passband; andthe tuning circuit is configured to tune the frequency response so thata signal bandwidth of the first RF QHRS and a signal bandwidth of thesecond RF QHRS are each within the passband.
 10. The tunable RF duplexerof claim 8 further comprising a tuning circuit wherein: the RF filtercircuit has a frequency response that defines a stopband; the RF filtercircuit is tunable to shift the stopband; and the tuning circuit isconfigured to tune the frequency response so that a signal bandwidth ofthe first RF QHTS and a signal bandwidth of the second RF QHTS are eachwithin the stopband.
 11. The tunable RF duplexer of claim 8 furthercomprising a tuning circuit wherein: the RF filter circuit has afrequency response that defines a passband and a stopband; the RF filtercircuit is tunable to shift the passband and shift the stopband; and thetuning circuit is configured to tune the frequency response so as toshift the passband so that a signal bandwidth of the first RF QHRS and asignal bandwidth of the second RF QHRS are each within the passband, andto shift the stopband so that a signal bandwidth of the first RF QHTSand a signal bandwidth of the second RF QHTS are each within thestopband.
 12. The tunable RF duplexer of claim 8 wherein the RF filtercircuit comprises a first RF filter and a second RF filter and wherein:the first RF filter is connected between the first hybrid coupler andthe second hybrid coupler and is operable to reflect the first RF QHTSback to the first hybrid coupler and to pass the second RF QHRS to thesecond hybrid coupler; and the second RF filter is connected between thefirst hybrid coupler and the second hybrid coupler and is operable toreflect the second RF QHTS back to the first hybrid coupler and pass thefirst RF QHRS to the second hybrid coupler.
 13. The tunable RF duplexerof claim 12 wherein the first RF filter comprises: a series LC resonatorcoupled in shunt between the first hybrid coupler and the second hybridcoupler, the series LC resonator having a series resonant frequency; anda variable capacitive element having a variable capacitance, thevariable capacitive element being coupled in shunt between the firsthybrid coupler and the second hybrid coupler such that the series LCresonator and the variable capacitive element have a parallel resonantfrequency.
 14. The tunable RF duplexer of claim 13 wherein the second RFfilter is identical to the first RF filter.
 15. The tunable RF duplexerof claim 12 wherein the first RF filter comprises a plurality of SurfaceAcoustic Wave (SAW) resonators, wherein one of the SAW resonators isconfigured to reflect the second RF QHTS back to the first hybridcoupler and to pass the first RF QHRS to the second hybrid coupler. 16.A radio frequency (RF) transceiver having a distributed duplex filteringtopology comprising: a tunable RF duplexer configured to input an RFreceive input signal that operates within an RF receive band of an RFcommunication band from an antenna, generate an RF receive output signalthat operates within the RF receive band in response to the RF receiveinput signal from the antenna, and simultaneously output an RFtransmission output signal that operates within an RF transmission bandof the RF communication band to the antenna and input the RF receiveinput signal from the antenna; a low noise amplifier (LNA) comprising alinear amplification stage and a tunable high-selectivity bandpassfilter, wherein: the linear amplification stage is coupled to receivethe RF receive output signal from the tunable RF duplexer, and isconfigured to amplify the RF receive output signal; and the tunablehigh-selectivity bandpass filter is tunable to provide a passband withinthe RF receive band, and is coupled to receive the RF receive outputsignal after amplification by the linear amplification stage.
 17. The RFtransceiver of claim 16 wherein the LNA further comprises atranslational filter, the translational filter being coupled to receivethe RF receive output signal that is output from the tunablehigh-selectivity bandpass filter.
 18. The RF transceiver of claim 17wherein the translational filter comprises an intermediate frequency(IF) filter, wherein the translational filter is configured to translatea frequency response of the intermediate frequency (IF) filter having apassband at an IF into an RF frequency response with a translatedpassband within the RF receive band.
 19. The RF transceiver of claim 18wherein the translational filter further comprises an RF passive mixerconfigured to mix the RF receive output signal with a local oscillatorsignal.
 20. The RF transceiver of claim 16 wherein the linearamplification stage is configured to receive the RF receive outputsignal as a single-ended signal and output the RF receive output signalas a differential signal.